This is a post that I started many months ago. Now that I am winding up my restoration activities it is time to post it, complete or not:
I probably paid too much for this, but it did come with the manual and these machines appear to be rare and most definitely have a place in history, for it seems likely that Hughes beat Tektronix to the market with a storage oscilloscope. The seller did not know anything about it, sadly it had been pillaged of its 8 12AU7s and 2 12AX7s. (I am giving the seller the benefit of the doubt here.) Fortunately, I had enough strong 12AU7s and 12AX7s on hand to replace them. This example has serial number 635.
It is quite nicely made but not even close to Tektronix standards. The bandwidth is a very limited 500kHz* and the timebase triggering is fussy at best. The timebase does not have a trigger shaper (usually a Schmitt trigger followed by a differentiation CR), this probably accounts for the general fussiness of the timebase. The Tektronix 564 storage scope came out soon after and offered 10MHz, a proper timebase, dual timebases if desired, not to mention a wide range of X and Y plug-ins. It was also usefully smaller. Speaking of the Hughes manual, it is awful, there are several circuit and description contradictions making it difficult to follow the (extremely limited) calibration directions. At the age of 56, even with diopter 2 reading glasses, I need a magnifying glass to read the schematics.
* The WB/4 Y plug in limits this further to 250KHz.
The ESSENTIAL thing with these CRTs is to NEVER place them face down. I don’t yet know on 1/30/2014, if this one is OK.
Here, I provide a description of the principal of operation of the Memotron tube, that I have stolen from the manual and re-written somewhat to hopefully make things either more clear or less muddy.
The Memotron tube has two electron guns, a writing gun and a flood gun.
There is a dielectric storage mesh placed before the viewing screen. The flood gun sprays the dielectric storage mesh with a (supposedly) uniform barrage of low-velocity electrons leaving the dielectric surface at the flood gun cathode potential. The high-velocity electron beam from the writing gun charges regions of the storage surface positive by secondary emission*, thus creating areas which are partially transparent to the flood electrons. The flood electrons which pass through are accelerated to high velocity producing a continuously visible image of the positive electrical charge pattern stored on the dielectric surface: as long as the flood gun is emitting, the positive and negative potentials of the charged pattern on the storage surface are maintained thus sustaining the image.
A second mesh called the collector mesh interposed between the writing gun and the storage surface is used to discharge the charge pattern on the storage surface the image by briefly lowering the voltage of the collector mesh.
This technology is sometimes referred to as bistable because a given area on the storage mesh will either be at flood gun cathode or collector mesh potential.
*Bombarding electrons can impart enough energy to surface electrons to cause them to break free, electrons breaking free in this way is known as secondary emission.
* 2/2/2014, well I know now that the CRT does store but I suspect it is not in its prime! The contrast is poor and the background brightness has to be quite high before the trace can be seen, more on this in the design notes below. I knocked up a “pinger” consisting of a 3H choke in parallel with a 1µF cap in series with a 10k resistor. I placed about 20Vdc across the circuit to store energy then quickly disconnected it resulting in the following damped oscillation (timebase is 20mS/div):
Contrast Enhancement Oscillator.
There is a blocking oscillator that applies +15V, 10µS positive pulses at a rate of 1000PPS to the storage mesh, supposedly to enhance contrast by decreasing the background brightness. Though the pulse amplitude is adjustable, adjusting it appears to make no difference at all. This is most likely due to the condition of the tube, however in truth, I have not made the effort to fully understand the action of the pulses on the CRT storage system.
This simply reduces the CRT collector potential to nearly zero at the same time as briefly stopping the contrast oscillator.
This is a simple DC coupled long-tailed pair with gain such that 9v pk-pk on the input terminals will produce full scale deflection of the CRT beam. The amplifier is served by a plug-in that in this case is a DC coupled differential pre-amplifier type WB/4. A dual channel chop/alt plug in WB/D1/11 was also available.
The sensitivity of the WB/4 is 10mV/div to 50V/div with a bandwidth of 250kHz. The legend on the front says “wideband”. Even during the era of this scope, 250KHz was not wideband! The chassis is mounted on rubber anti-vibration mounts however, the sensitivity of the unit does not seem to warrant such a detail. Perhaps this is more important when capturing the slow events that this scope is clearly intended for. I would think that it was used to observe mechanical systems, especially since an aircraft company (Hughes) put for the effort to create it. There must have been a critical need for it. It would be interesting to learn just what is was used for, maybe something, in the arrogance of hindsight, that we now regard as obvious.
This DC coupled amplifier is similar to the Y amplifier except the degree of coupling between the cathodes can be varied to control the gain. The gain is set to cause the sweep to occupy exactly 10 divisions across the screen, best done using a marker generator. The amplifier may accept an external sweep and in this mode the gain may be continously adjusted by 10:1, a further 10:1 being available on the Ext Sweep portion of the Time/Div switch. The sensitivity is 0.5 to 50V/div with a bandwidth of 250KHz.
The timebase is switched on and off (trace, flyback) using a bistable multi-vibrator which is pretty typical practise. However, there is no Schmitt trigger between the vertical signal and the control multi. There is a cathode coupled trigger amp thats allows triggering on positive or negative going signals. The DC output level of the trigger amp is variable so that the trigger signal may be shifted relative to the DC trigger point of the control multi, in this way the DC level on the signal at which the timebase starts may be set of varied. Because there is no fast squaring (Schmitt Action) of the triggering signal followed by differentiation to produce a sharp trigger pulse, this system is unsuited to fast events.
The timebase essentially consists of a bistable control multi and a positive going sweep generator. The multi is anode triggered with the triggering signal being applied via a diode (so that it is negative going) to the input tube that is off, awaiting a trigger. This forces the anode voltage down, this change is communicated to the grid of the on tube causing it to cease conducting and the multi to flip states and the off (input) tube turns on, the anode voltage falling further. The multi remains in this state for the duration of the sweep. The resulting negative pulse is used to turn the timing capacitor discharge tube off and the capacitor starts to charge. Linearity of the charge process is ensured by bootstrapping the charge to a control tube across the charge resistor, maintaining the voltage across the charge resistor constant and thus the charge current also. The sweep voltage rises approximately 90V and is communicated to the grid of the off tube of the multi, causing it to turn back on resulting in a positive going pulse at the grid of the sweep discharge tube, turning it on, and discharging the timing capacitor. With this action, the input tube of the multi is now returned to the off state and the circuit is ready to receive the next trigger. Delay between the discharge process and the next sweep to ensure that the discharge is complete is accomplished by an RC time constant on the grid of the discharge tube causing it to turn on a little after the multi flips back to the sweep state. This is a simplified hold-off, in better designs the hold-off time constant is switched to different values with the sweep range switch. There are 6 ranges, 10µS/div to 1S/div together with a 1X, 2X and 5X times multiplier such that the range of the timebase is 10µS/div to 10S/div. There is also a 10X continuous multiplier control.
This is a clever solution to the problem of coupling the unblanking signal (positive pulse from the timebase control multi) to the beam control (usually grid) element of the CRT. What Hughes did was to turn a self excited oscillator on and off using the unblank pulse. The oscillator runs at around 10.5MHz. The oscillation is fed via a low impedance transmission line to a receiving coil and capacitor that is tuned to the same frequency as the oscillator. The resulting voltage is doubler rectified and applied to the grid of the CRT. Neat.
This consists of a HV oscillator driving an air cored transformer at around 300KHz. A sample is taken from the rectified output and via an amplifier is used to control the screen voltage of the parallel oscillator tubes thereby regulating the output in the usual way. The PDA voltage is 4.3KV while the grid and cathode supply is around -2.85KV.
“Low” Voltage Supplies.
Per Tektronix (and everybody else after Tek showed the way), there is a precision -200V rail, this is used as the reference for stacked +200V and +450V rails. There is also an unregulated +325V rail.
The first job was sorting out replacements for the missing tubes and using Deoxit on all the tube sockets. Then came replacing burnt out power supply protection resistors, a couple of the caps required re-forming and most probably replacement soon. Once this was done, the regulated supplies came in a little low but all in tolerance, the drop-out point was well below line voltage. It was not long before I realised that something was not quite right with the HV (CRT) supply which is a 300KHz high frequency oscillator type with the oscillator amplitude being modulated by amplified DC error feedback from the output to the oscillator screen grid regulator. It uses 2 1X2B rectifiers and on checking them, the negative supply rectifier was very weak so I replaced it too. I was then able to get the negative (-2.85kV) and PDA (+4.3kV) supplies in tolerance though the regulation is not solid and the voltages do vary a little as the intensity control is operated. I can find no obvious reason for this, all components are good and the screen regulator amplifier tube is good too. Here are a couple of views of the HV PSU:
I particularly like the air-cored, pie-wound HV transformer and the intensity and focus control universal joints. The rectifier heaters are supplied by the loops around the base of the transformer.
Interestingly, while the PSU reservoir caps were OK (well, after reforming 2 of them anyway), I did find it necessary to replace the regulator output shunt caps. (Caps that are not designed to handle rectification charging currents are less robust, insulation-wise.) One of them was tucked at the lower back corner of the PCB with the power transformer on the foil side so I had to remove the power transformer from its mounting. If you click on this picture, you will just be able to make out the offending cap behind the wiring in the lower right-hand corner:
And here is the transformer side with the plug-in cage, the CRT and shield and the HV PSU removed:
If you click on the picture, you will be able to see that the power transformer is off its mountings and actually could be swung right out of the corner to reveal the foil pads for the offending capacitor. While you are there, look for the nice universal joints near the front panel that drive insulating rods that connect to second universal joints on the intensity and focus pots in the HV PSU.
Having decided to dispose of my collection, this is the first item I put on ebay and it sold in the first day, clearly I priced it too low at $135 however, it is the first thing I have ever sold on ebay, live and learn!
My ebay name is triodeguy.
I have many items that have not yet made it to the blog and as I sell them I will post the description and pictures so as at complete this documentation project, albeit without the technical and restoration narrative.
Here is the description I used on ebay:
GE regulated power supply used to power a transmitter modulator. Who knows, you may have listened to your favorite station when powered by this unit!
I fitted a terminal block to the cut wire ends. The wire nut connects the line to the B+ transformer to allow the B+ to be remotely switched. A B+ time delay could be installed here.
The raw B+ supply is configured as choke input, given that the massive 10H choke is rated for 340mA and that there are 3 6AS7/6080 series tubes, I would expect this power supply to be conservatively rated for 300mA.
The rotary switch allows the meter to indicate total current, current through each section of each series tube or output voltage.
I have powered it up and it delivers 300V DC that can be varied from 255V to 415V.
I load tested from no load to 300mA (at 300V) and the voltage did not shift more than a volt however, it IS old and I will not guarantee your satisfacton.
This is pro gear and has just one electrolytic capacitor, the rest are paper in oil.
Use vintage the pictures to form your own impression of the condition of this unit, NOTE, while testing it, I did find one dead 6AS7 section so I replaced that tube with a good 6080.
Again, being vintage, I DO NOT guarantee proper operation so this unit is sold as is.
It weighs 58 lb and measures 13″ x 10.5″ x 18.5″
And here are the pictures I used:
I have decided that it is time to start letting go of stuff.
If you are interested in acquiring any of the items I have written about, please comment on this post and I will get back to you.
8/9/14, UPDATE: I took and passed the general class today. Now the learning (leaning) curve begins……
One area of vacuum tube technology that I have not (yet) explored is (amateur) radio. I have a technician license and have been putting off doing anything more because I found the antenna (and ham jargon) issue daunting. One thing is that I want the feedpoint at the house, not in the middle of the yard! This makes a classic dipole less attractive given my house and yard. A friend (ham mentors are called Elmers) suggested a loop and sent me an excellent article describing the setup and performance of a loop antenna. Because it is a loop, the feedpoint can be anywhere so I have mine where I want it! The loop length is made equal to the wavelength of the lowest frequency of interest, in this case 3.7MHz giving 272ft. The larger the area enclosed by the loop the better, so the ideal shape is a circle. Mine is a pentagon because that gives me the best compromise given the wire length and the trees I have. The feed is RG8X 50Ω coax. I now need to learn how to use it!
I have a Heathkit SB-102 transceiver that I have repaired and checked out into a dummy load. It has a solid state Variable Frequency Oscillator (VFO) that is supposed to be significantly more stable than a tubed VFO. Using a MFJ Deluxe Versa Tuner II with the Heathkit, I obtained Standing Wave Ratio (SWR) figures of 80M, 1 to 1.1; 40M 1.2 to 1.3, 20M, 1 to 1.1 and 10M 2.5 at best. I need to upgrade my license so that I can use voice (SSB) on bands other than 10M! Some people use a 2:1 balun with a loop and I do wonder if this would yield better performance at 10M? I am a beginner….
By the way, the J in MFJ may stand for junk. The vanes of the variable condensers in my unit were rubbing due to massive slop in the spindle bushings that allowed the vane assemblies to flop around. I bought the unit on ebay however, it shows no signs of use much less abuse, so I think this issue is due to rubbish quality parts. I was able to slip plastic strips into the spindle bushings and remedy the problem. Even so, I would not use it with solid state transmitter finals.
The wire is 14ga stranded and insulated house wire. I placed loops of rope around the trees / branches that looked to be good locations for suspension. The loops are fitted with pulleys. Each suspension point on the wire is fed through an insulator tied to another rope (1/4in Daycron) that passes through the pulley down to ground level. In this way, the suspension points can be dropped down to ground level when the wire snaps (it will for sure). The length combined with the elasticity of the suspension ropes provides some accommodation for tree sway however, the mid-point suspension has a dead weight load consisting of two house bricks. (The dead weight tension method is used for the suspension of railway power catenaries.) At the lowest point, the chimney it is 24′ high and at the highest, around 40′. I hope this arrangement will accommodate most wind activity! The feed point is tied to the chimney of my house (which seems a better scheme than tying it to my neighbours house). It consists of a small plastic Radio Shack project box fitted with an eye bolt for the anchor. The ends of the antenna come in, then out and back in again through small holes providing sufficient grip to prevent them pulling out. The feed cable comes in through the bottom of the box and a tie wrap inside prevents it pulling out. All the entry points are sealed with silicon goop.
Oh, I had great fun playing with a potato gun to launch 3 of the suspension points……….
I have decided to take on writing up at least some of the GR kit in this collection. This unit was for sale at an attractive price given that it included a very neat GR 1201-B unit regulated power supply. It did not work properly, ceasing to pulse whenever I turned the pulse width control much either side of centre. I did not do well in diagnosing the problem, diving in too deep before simply studying the thing. The result was a cycle of picking it up and putting it aside. A very perplexing issue was that the pulse width was off by a factor of ten and I got into all sorts of trouble with that one, sure that it was acting as a frequency divider even though that made no sense. Finally I saw it, a previous muddler had (re)connected the pulse width pot incorrectly, damn! And so it now works quite well. The Pulse Repetition Frequency was low and again, manual in hand went through all sorts of component value conniptions (based on the manual) only to realise upon studying the circuit calmly, that the tube condition would affect the PRF. The manual makes no mention of this yet looking at the circuit it must be, and is. I simply went through a number of tubes until the PRF came in correct at the calibrated position of the PRF pot. Such dependence on the tube is not necessary and is surprising to me considering the reputation GR has.
I don’t have a lot more to say about it. It works well after many the self-made detours. The manual claims that it can deliver rise times of less than 18nS and fall times less than 10nS. I was able to see 18nS and 10nS on my Tek 475 (250MHz bandwidth) but not less. It does find quite a bit of use on my bench.
Repetition Rate, 2.5Hz to 500kHz with calibrated points in 1-3 sequence from 10Hz to 300kHz plus 500kHz. Continuous coverage with uncalibrated control.
Duration, 100nS to 1S in 7 decade ranges.
Pulse Output Levels: + and – 40mA pulses, each 40Vpk into internal 1k load. DC coupled with DC component negative wrt ground.
The 1201-B power supply provides a regulated floating 300VDC up to 70mA and 6.3VAC at 4A. In the pulse generator, the floating voltage is referenced to ground at -150V and +150V.
The pulse repetition generator is a Schmitt trigger with a RC circuit charge/discharge appended. The C of the time constant is charged from the plate circuit of the A section via the R of the time constant that is between the plate and the grid, thus the charge and discharge of the grid drives the A section. If the A section is off (plate high), the capacitor will charge through the R until the A section turns on*, turning the “B” section off. This action happens suddenly due to the regenerative switching action of a Schmitt trigger. The B section has an inductor in its plate circuit and the sudden release of energy due to the B section turning off produces a sharp pulse that is used to trigger the output pulse timing circuit. When the A section turns on, its plate voltage will fall and the capacitor will discharge until the A section turns off again. In this way, the oscillation swings takes place within the hysteresis of the trigger. The reason for PRF dependency on the tube as well as the RC values is that the hysteresis of the circuit will vary from tube to tube. The PRF ranges are controlled by switching in different values of C and continuous control is provided by making R variable from the front panel.
* The manual incorrectly states this as off. Here is the repetition generator circuit:
V101 serves as a current source when the circuit is in generate mode (switch position O), allowing the Schmitt trigger to freely swing around its hysteresis point. In position A, it acts as a driver for the Schmitt trigger to allow the circuit to be driven from an external source.
The positive trigger pulse from the repetition generator is applied to the pulse timing circuit and then the whole bloody thing goes belly up, or at least it did until I spotted the wiring blunder.
Q101 is normally on and with it, V103A and diode V103B, holding the voltage at the junction of R118 & R122 at a level determined by the setting of R125, the Pulse Duration control. Since Q101 is on, V105 is also on producing a current in the positive pulse output load resistor, R130.
A positive pulse trigger pulse from V102 (the pulse repetition generator Schmitt circuit) turns Q101 off and with it, V105 and V103A. Q102 turns on, and with it V106, producing a current in the negative pulse output load, R133. C2 begins to charge (ramp up) via R118 and the grid voltage of V104 rises until this tube conducts and the Schmitt circuit V104A and B changes state, sending a positive triggering pulse to Q102, turning Q102 off and Q101 on, re-establishing the initial circuit conditions. The circuit is now ready for the next cycle initiation pulse from the Pulse Repetition Generator. The pulse duration is therefore controlled by the time constant R118, C2 and the initial potential on C2 established by the Pulse Duration control.
The pulse output circuit consists of pentode V105 and V106 that are switched as described above by the transistor bi-stable, Q101 and Q 102. In the initial state, Q101 is off and with it, V105 causing the output pulse to rise to ground, going low when the circuit changes state. V106 operates in the opposite direction. The screen voltages set the zero bias currents at 40 to 45mA and since the output loads are 1k, the pulse amplitudes are 40 to 45 volts in maximum amplitude.
I have one of these neat power supplies each for my 1217-B Pulse Generator and 1210-C RC Oscillator so I thought I would do a quick post on it. It supplies a floating precision 300V at 70mA and 6.3V at 4A.
A search on the net did not bring up the schematic for this unit so I have traced it out:
(P.S. After doing this I did find a link to the manual for this unit! Go here)
Apart from the primary control loop from the output to the series pass tube via the error comparator (12AT7), amplifier (6AN8 P) and cathode follower (6AN8 T), there are two other signal paths: C531 and R533 feed input fluctuations forward to the grid of the series pass tube in cancelation phase via the amplifier (6AN8 P) and R540 serves to make the open loop gain infinite thereby allowing very low output impedance. The technique of regenerative feedback to allow infinite open loop gain coupled with degenerative feedback to realise extremely solid stability and freedom from drift is also used in the TS-375 A/U VTVM DC amplifier which is the subject of my previous post.
I haven’t characterised the impedance vs frequency however, the use of a grounded grid pentode amplifier with a cathode follower to drive the grid of the series pass tube clearly indicates that it was designed to have excellent performance at high frequencies. My own regulator designs usually include feed-forward (DC coupled) but I have never attempted to make the open loop gain infinite.
Here is the inside with the PCB flipped up:
INTRODUCTION: This unit potentially has practical value to me since it is both an AC and a DC Valve-Voltmeter, also as an AC meter at 100MHz plus, it has a much wider bandwidth than my AC only HP 400H (4MHz). (By the way, Valve Voltmeter because I am of English origin and I think that valves are SO much more interesting than toobs, you see.) I have had this VVM for quite a while and it was one of those projects that I flirted with and then set aside, again and again. The reasons for this were 1, it did not work properly and 2, I could not understand the circuit; I finally re-drew the DC amplifier and meter circuit for clarity and I have included this diagram in the theory section below.
CONDITION: It is in excellent condition with tubes that are effectively free from microphonics and balance perfectly, and the balance once set, stays put. However, I could never get it to read accurately and consistently. I had removed the meter cover and gently blown on the needle, it seemed completely free however, I finally realised that it was in fact sticky. The disturbing force due to blowing is quite a bit stronger than that due to the available magnetomotive force! Since the meter is quite well sealed, I felt that the problem was not likely to be particles in the gap so I gingerly backed off the top pivot. Sure enough, the meter cleared and now does move under electrical stimulus freely and can be reversed at any point without sticking. I cannot offer any explanation as to why the pivots were dragging other than perhaps corrosion. Having said that, I do wonder at the quality of the (Simpson) movement; The mass balance of the needle is very poor despite the presence of balance weights, and it can only be calibrated and used either flat or vertical. I set it up flat because I will use it on a bench, not sitting on a shelf. The picture below shows the large mechanical zero error due to the mass unbalance:
DESCRIPTION: It is an extremely neat unit, and typically for military equipment is enclosed in a grey aluminium case. (By the way, I do have the knobs from the terminals, I simply forgot to replace them before taking the picture.) Both the AC and DC probes are present however, other than the power cord and spares fuses and lamps, none of the other clips and cords are present. Here it is showing the probe compartments open:
The AC input at the terminals is routed via the rectifier probe which is stored on a mounting clip that connects to the probe tip and probe grounding ring. To make accurate measurements above 100MHz, it is necessary to connect the source signal ground directly to the probe ground ring and test tip directly. Even at 40MHz, the leads should not be longer than 3 or 4 inches, according to the manual. The probe tip and ground ring with their respective cradle contacts are both shown in the picture below. The ground wire should not be more than at the most 4″ long. This means that the source ground and test points must be close together. One problem was that the tip of the AC probe was shorting onto the case, you can see where the paint has been scratched. I managed to adjust the cradle a bit but it is still close:
Here is the spares compartment:
And here is what is contains:
And finally, before we get in deeper, here is the inside, note the two spare knobs and spare rectifier on the left by the pots:
AC and DC voltage ranges, 1.2, 3, 12, 30, 120, DC only, 300.
Input loading, DC 30MΩ, that is useful for me!
AC, 5MΩ shunted by 70pF at the panel terminals or 5pF at the AC probe tip. The manual has a graph showing that the resistive component of the loading falls (the load increases) with frequency, two points from the graph being 5MΩ at 10KHz and 90,000KΩ at 100MHz. A good scope with a top quality X10 probe can do better, typically 10M shunted by 2pF. However, when this unit was produced, scopes that could match the bandwidth of this meter existed only in engineer’s dreams! (Engineers are weird like that, I KNOW!) Also, this unit is much more handy than a lab grade scope of the era in any case.
DC, all ranges 3% FSD.
AC, 10 to 50Hz, 5% with correction curve
50Hz to 100MHz, 4% without correction
50MHz to 150MHz, 3% with correction curve
150MHz to 300MHz, 8% with correction curve
The meter meets the % FSD specification on the AC and DC ranges. I made at least 3 spot checks on each range after some tweaking. It is worth noting that as is normal for meters, the specification is given as % FSD; The % of actual values is often quite large but within 5% except for the actual value at 1V on the 3VDC range which is -6.25%, this translates to -2.08% FSD. Most DC errors were on the low side which suggests that I might be able to set the calibration better. Having said that, the AC errors were a mix of high and low so to actually accomplish better calibration would be very tricky and most likely not stable, so I will not attempt to “improve” it.
It is essential to carefully zero the meter with the input terminals shorted on each AC range, you cannot simply switch from range to range. The zero variation on the DC ranges is negligible. This is very likely due to diode contact potential about which I say more in the theory section below.
I checked the frequency response. It is claimed to be flat over the range 100Hz to 100MHz which if true is excellent, for the period of this meter at least. I checked this aspect out using a HP 8601A generator terminated into a 50Ω through BNC with the AC probe and ground ring connected right at the termination, on the 3VAC range I observed a 1dB increase up to 97MHz increasing to +1.5dB at 110 MHz which is the generator limit. Clearly there is a resonance somewhere in the signal path yet I am impressed. This meter is better than it seemed to me at first, second and third blush. This is the connection arrangement to the signal generator:
Overall, this seems to be a capable instrument, at least in the context of its vintage. However, it is essential confirm to that it is reading accurately in the range you desire to use by connecting a known accurate meter in parallel with it and testing it using a DC or true sine AC supply. This is not as silly as it sounds because the point of this meter is the extremely light load it applies to the circuit under test and its wide bandwidth. Your DVM probably cannot match it in these aspects.
THEORY OF OPERATION:
Essentially, it is a DC amplifier and meter. A good DC amplifier is not a trivial proposition because of drift. Most VVMs rely on large amounts of degenerative feedback to stabilize the circuit, however this can result in problems with HF oscillation. Perhaps that is why the HP 400H mentioned above is limited to 4MHz. This design relies on both regenerative and degenerative feedback that I will attempt to explain with the aid of the manual. (The HP 425 Micro-Volt/Ammeter uses a light chopper that produces a rough squarewave that is proportional to the DC current and in this way, completely removes drift. This is followed by an amplifier having a passband at the chop frequency that suppresses other frequencies (including 60Hz) heavily. In fact the chop frequency is chosen not to be harmonically related to line frequency to further ensure no hum on the extremely fragile signals. The amplified chop is then demodulated, rectified and applied to the meter. I think this exemplifies that DC amplifiers are truly non-trivial!)
Because it is a DC device, AC currents must be rectified and this takes place in the AC probe. The result is a peak voltage and this is corrected such that the resulting DC voltage presented to the DC amplifier is equal to the RMS voltage, if the AC current is truly sinusoidal*. This correction is applied by R-111 in TS-375/U and fixed R-133 plus variable R-134 in TS-375A/U. If the waveform is other than sinusoidal, a form or crest factor must be applied. The AC rectifier is right at the probe tip so that the AC signal does not have to travel along a wire and this is how the wide bandwidth is realised.
* The GR 1201-C oscillator that I was using to test and calibrate the AC ranges with has a slight clip on the positive peak (I don’t like GR equipment other than their bridges, heresy I know) and this was sufficient distortion to confound my efforts. The HP DVM I used as a reference would indicate the true RMS of this signal but the VVM would not. I ended up using a HP 205AG that produces a clean waveform with enough available amplitude to test the 120V range.
First here is the simplified circuit:
The screen cross-coupling provides the regenerative paths while the DC plate to grid offsets, (coupling batteries in the diagram) provide the degenerative paths and the output terminals. The plate resistors and the tubes comprise the 4 arms of a bridge and so you can think of the meter as being across the bridge from one plate/plate resistor node to the other plate/plate resistor node, via the coupling batteries. This visualisation may help you to see the bridge.
Degenerative feedback brings stability, any error is returned to the input as a cancellation voltage. However there is a limit to the degree of degenerative feedback possible with a conventional amplifier due to the limits of the single stage amplification. In this design, the additional gain needed for stability is obtained by regeneration, permitting a higher level of degenerative feedback and thus higher voltage stability in this single stage circuit than would otherwise be possible. Quoting from the manual “By this method a gain ratio greater than the actual gain of the amplifier tubes proper can be realized.”
Now to the actual circuit: The “batteries” have been replaced by gas voltage stabiliser tubes, the internal resistance of the tubes being low enough to act effectively as batteries in this circuit. The strike currents for the stabiliser tubes is provided by the keep-alive resistors, R-106 and R-107, each being 100k. These terminate at the negative supply lead. The keep alive currents are too great for each plate circuit so the positive end of the strike tubes are fed via V-103 cathode followers from the + supply lead, the cathode followers being driven by each screen grid. In this way, the practical circuit emulates the simplified circuit. The resistor R-108 is included in the cathode circuit to allow for the voltage dropped across R-106 and R-107.
I found the circuit as drawn difficult to follow so I re-drew it thus:
The regenerative (positive) feedback paths are obvious, R-104 limits the regeneration to a prescribed level. What was not obvious to me was the degenerative (negative) paths and the meter circuit so my re-draw show how the meter is connected to each plate/plate resistor node of the bridge via the screen CFs and the gas tubes. The gas tube keep alive resistors are effectively in parallel (Thevenin) with the plate load resistors. The degenerative feedback paths are shown in blue and red. Taking the blue first, the path is plate to ground then via the input voltage, to the input grid. Since the whole circuit is floating, the ground connection does not affect the operation. The red feedback path includes the meter which has a negligible resistance of 1K. Any grid current is orders of magnitude too small to cause sensible deflections of the meter which is driven by the current due to the unbalancing of the plate voltages.
(At this point a question arose for me: Why isn’t the meter simply connected directly across the plates? The answer is that the resistance value of the meter path would reduce the regenerative feedback too much. As it is, the CFs isolate the R-104 regeneration path from the meter path. It is a clever circuit.)
At the point of optimum adjustment, the screen-grid circuit is critically regenerated and on the point of self-sustaining oscillation if the degenerative paths to the control grids were disconnected. The manual suggests thinking of the circuit as an amplifier having infinite gain due to the critical regeneration which is completely degenerated by external feedback paths (the blue and red paths); thus the stability improvement possible due to degenerative feedback around any amplifier is in this case, carried to the limit.
At this point of critical regeneration, the screen grids do the actual work of unbalancing the bridge to produce an output (proportional to the input) and the control grids simply serve to initiate the unbalancing action. For any steady value of input, the potential of each control grid with respect to its cathode is the same and the potential difference across the grids is zero. The value of zero grid voltage excursion is that it entirely removes the effect of the curved grid voltage, plate current relationship and the amplifier is strictly linear in its input to output relationship. As I said, it is a clever circuit!
I mentioned diode contact potential earlier: A hot cathode diode will develop a negative potential on the plate with no ac input, this potential is termed the “contact potential” of the plate wrt the cathode. The diode in the probe is compensated by adding a similar diode to the red side grid circuit. I suspect that the very slight change of DC resistance across the compensating diode as the range is changed may account for the need to re-zero the balance every time the ac range is changed.