INTRODUCTION. I had one of these when I was a kid. It was my first experience of a ‘proper’ oscilloscope, capable of making measurements and started my (admittedly eccentric) fascination with these machines. This was around 1974 and my dad, a Chartered Electronics Engineer and Senior Scientific Officer with the UK Met Office Cloud Physics lab, still used a Cossor 1035 MKIII at his development bench while the technicians were using Tek scopes. The original version when it was introduced, may have been the first dual-beam, triggered oscilloscope capable of making measurements. Since I wrote that, I found a June 1950 Wireless World advert for the original 1035 and from the picture I can see that it did not have the timebase delay that the MKIII has. Tektronix did not come out with a dual channel scope until the 530/535 in 1954 so Cossor did beat them to it. However, once Tek did bring the 530 series, everything else was in the shade. That it took HP 10 years to catch up (barely) really says it all for Tektronix. Anyway, here is the Cossor ad:
ARRIVAL. I had been wanting one of these and a guy in England contacted me saying he had just acquired one and was looking to sell it. The pictures showed a MKIII example in good condition so I took a chance on it and amazingly, just 5 days later, UPS arrived saying “we have a machine for you”. I unwrapped it with no little anticipation and was much relieved to find it undamaged. I think it is handsome, it is also surprisingly capable:
SPLIT BEAM CRT. It is based on what as far as I know is a unique Cossor technology, the split-beam CRT. (A. C. Cossor got into business initially in 1859 as a manufacturer of scientific glassware, later X-ray tubes then vacuum tubes, diversifying into electronics after WW1.) The tube uses single Y plates that are placed either side of a beam splitter that is in mechanical and electrical contact with the final anode. The location of the Y plates either side of the splitter largely shields each beam from ‘fringing’ or interference by the opposite Y plate. The shielding is rendered perfect at the anode end by the contact with the final anode. At the screen end, fringing is reduced by means of two ‘bucking wires’ that are cross-connected to the opposite deflector plates and serve to annul the stray fields such that the interaction between beams is reduced to less than 1%. Correction for equal beam brightness is made using a small permanent magnet near the socket end of the tube that by deflecting the beam before it meets the splitter plate can be adjusted to obtain equal beam brightness. (Source: The Cathode Ray Tube And Its Applications, Parr and Davie.)
I recall that the unit that I had as a kid exhibited the best beam definition I had ever known and this seems to be true, even the best of the Tek CRTs falls a little short and most, far short. It is simply beautiful to use.
CONDITION & REPAIR. Upon arrival, I found it to be fairly dirty inside, otherwise in excellent condition. I brushed the loose dirt away, I may wash it in the summer when there is strong sun to dry it. Then came the usually Deoxit treament of all the tube sockets. Next was to replacing the dried reservoir and smoothing caps (I disconnected them and checked them using a bench PSU). The flyback suppression coupling cap (visconol) was leaking oil so I replaced it right away. On turning it on I was rewarded with traces but a whole lot of hum showing. This turned out to be due to leaky coupling caps in the A1 amp causing heavy current draw that was pulling the B+ down hard enough to cause the regulator to drop out. Replacing those resulted in quiet traces and we were on our way! I wound up replacing most of the small paper in oil caps including the Plessey “toffee” types used for the timing of the timebase and delay circuits. They were all leaky causing extremely poor linearity:
The mains selector permits operation on 120 V so I fitted a new US style power cord. The only tube replacement required was the short suppressor base (in the US, dual-control) pentode used in the timebase. (These types have an unusually close wound suppressor grid to achieve the short base.) Cossor used a 6F33 which I replaced with a CV2209 (which I think may have been a special quality tube type for Racal). After taking care of these simple issues, I was able to calibrate it.
Here it is displaying 1 mS markers and a squarewave that is synched to the markers:
Note the great trace definition! The distortion at the start is due to a disadvantage of the transitron based timebase, the Miller step, whereby the anode voltage drops a small amount suddenly at the initiation of the sweep. I say more about this under the design notes below.
PERFORMANCE & HISTORICAL CONTEXT. It has two AC coupled channels, the A1 channel having a response of 5 Hz to 5 MHz with a maximum sensitivity of 10 mV/cm; the A2 channel has a response of 5 Hz to 250 KHz with a maximum sensitivity of 1 V/cm.
The timebase has 9 ranges, 100 mS to 10 µS. X expansion is available permitting a maximum spot velocity of 250 nS/cm. The reason the timebase ranges are stated as time not velocity is because the time ranges relate to a vernier scale on the X-shift; time is measured by setting the vernier at zero with the start of the event at a chosen location on the screen graticule, then using the vernier, sliding the end of the event to the same location. The time is given by multiplying the vernier reading by the time range is use. Voltage on the Y axis is measured using the same slide-vernier technique, the value of the event height read from the vernier is multiplied by the Y attenuator setting value in use to give the voltage. This method seems archaic but in use is very simple and easy. In fact, the timebase delay technology introduced by Tektronix handles magnified event measurement in the same way. At that time, nobody had introduced regulated CRT HV supplies and this was a good way to avoid the effect of HV voltage related CRT sensitivity variation on measurement accuracy. (Again Tektronix introduced CRT supply regulation.)
Timebase delay is also included to permit event expansion, such as the viewing and measurement of the whole of a pulse across the screen width. The specification states that it will measure time and voltage to better than 10% which nowadays in our Parts Per Million world sounds hopeless, but in those days of oscilloscopes that were not capable of calibration was good news! In my estimation I would say it is better than 5%, not far short of the new standard set by Tektronix. I don’t know when the first 1035 MKI model was released but it must have been around or maybe slightly before the ground-breaking Tektronix scopes first became available. Unlike the first Tektronix, the 511, this one has regulated power supplies, absolutely unheard of before then. Tektronix did not regulate the power supplies in their scopes until the 511A.
The timebase is a triggerable sweep duration rather than frequency design and this permits time measurement without resorting to a marker generator. It can be operated in triggered or repetitive modes. Here is the first reference that I have found to this design again, from Parr and Davies:
Martin on the british Vintage Test Gear & Workshop Equipment forum pointed out the the EAC91 specified is a single diode triode, it looks like an error and may have been intended to be EBC91, thanks Martin!
WHAT IT LOOKS LIKE INSIDE. Before I get into a more detailed technical description, here are pictures of it with the covers off, there is a lot of space due to the original design having used octal tubes!
The A1 amplifier is on the shelf while the A2 amplifier is on the deck at the front. The power supply regulators are on the deck at the back.
The timebase occupies the front section with the CRT circuit at the back.
Note the handsome potted double C core power transformer! I also like the Colvern ganged wirewound precision potentiometers used for the measurement vernier scales.
Cathode Ray Tube. The design is based on a Cossor split beam, flat face CRT, the major consequence of which is that the Y amplifiers (referred to in the manual as A1 and A2) are single ended. This might sound like a major compromise and in comparison with more modern designs I suppose it is however, in use, the amps seem to provide highly accurate, symmetrical traces. Despite the lack of common mode rejection or DC heater supplies the traces are perfectly free from visible noise. The X and Y deflection plates are available for direct connection via slide doors in the oscilloscope cabinet.
The cathode is operating at -2kV with the first and final anodes at ground potential. A Post Deflection Acceleration anode is provided, normally run at +350V with provision to increase the PDA to 2kV when extra brightness is required. All deflection plates are run at DC ground potential to minimise trace distortion. The focus is exceptional despite the lack of astigmatism correction.
Here is another picture of it showing gate and sweep waveforms from a Tek scope:
Power Supplies. The basis of the power supply is a beautiful potted C core transformer. A single 330-0-330 winding serves both the positive and the negative rails, the positive rectifier being a tube and the negative side is handled by two selenium rectifiers.
The positive side is LC smoothed to provide a +350 V rail that feeds a classic series regulator having a triode connected pentode for the series pass element and a pentode error amplifier. The regulator supplies +200 V to most of the circuits in the scope.
The negative side is RC smoothed to a -300 V rail and a gas tube stabilised -150V rail that also serves as the reference for the +200 V regulator.
The CRT supplies are taken from a single HV winding and recified using selenium sticks both ways to provide plus and minus 2 kV, these supplies are RC smoothed.
Making Measurements. At this point, it is appropriate to further expound on the vernier-shift measurement technique used for both time and voltage measurements on this oscilloscope. Here is a diagram showing the electrical arrangement:
The method of measurement is; set the cursor of the appropriate vernier to zero on the scale on the panel, then use the shift potentiometer to set the lower (voltage measurements) or left (time measurements) part of the event to a chosen location on the graticule, usually the centre. Then operate the vernier until the other end of the event is at the chosen location. Read the value on vernier scale under the cursor and then multiply by the voltage or time range setting to give the voltage or time value.
The two variable resistors RV13 & RV15 are a ganged pair that comprise the shift control, the wiper of RV13 is connected to the vernier potentiometer RV14, while the wiper of RV15 is connected to the other end of the vernier potentiometer, the shift potential being taken from the wiper of RV14. RV13 & RV15 have no scale, they allow the beam to be shifted by moving the entire vernier pot up and down electrically whilst maintaining a constant potential across the vernier pot and thus, the calibration of the vernier scale. The vernier-shift network is fed from regulated supplies to maintain calibration.
The calibration is accomplished using RV12 whereby changing the setting of RV12 will change the potential across the network and thus the vernier pot, changing the amount the beam moves for a given movement of the vernier pot. RV12 is manipulated until the vernier scale accurately corresponds with a known voltage or time being displayed. For example, if the attenuator is set at 5 and a 5 V p-p event is on the screen, with the upper peak on the centre and the cursor at zero, then shifting it downward from zero using the vernier until the lower peak is at the centre should place the cursor over 5 on the scale.
The graticule is engraved on both sides that in combination with the flat screen makes it easy to avoid parallax error. It should be noted that due to the Miller Step, Cossor recommended avoiding the first 10% of the trace when making time measurements. Physically, the vernier pot is mounted at the front of the shift pair and operated by a spindle that runs coaxially around the spindle of the shift pair control spindle and has a nice hairline cursor knob over a vernier scale that is printed on the front panel. The spindle of the shift pair emerges at a simple small control knob in the centre of the vernier.
Here is what the controls look like:
The beautiful Colvern wirewound pots used for these controls are sufficiently linear that the use of a printed scale works very well. This is in marked contrast to the silly printed scales on the front panel of toy oscilloscopes!
Y Amplifiers. The A1 amplifier is the primary Y channel and the trigger / synch signal is taken from this channel. One aspect that is odd is that the A1 deflection is positive down, I would love to know the design rationale for this arrangement…….. This could have been overcome by mounting the CRT the other way up! The A2 amplifier is a simple differential pair and phase reversal is provided on the range switch so if the CRT had been mounted the other way up, it would have been simple to have the panel phase marking for the A2 channel correct. (I am actually considering this modification, if the CRT wiring will permit. All that is needed to correct the A2 channel (to maintain the correct front panel phase markings) is to reverse the grid connections from the attenuator to the opposite tubes!
Both channels are AC coupled which is a significant downside, probably necessary to avoid DC drift in the single ended amplifiers that would be hard to eliminate despite the use of regulated power supplies.
The A1 amplifier has a maximum gain of 3000 (69.5dB). It uses three RC connected pentodes and what the manual describes as “heavy negative feedback”. Each stage has adjustable bias, the bias and all plates and screens are fed from the regulated supplies except the final stage plate which is fed from the LC smoothed 350V supply. Seven switched HF compensated attenuation levels provide sensitivities from 10 mV/cm to 10 V/cm. The frequency response on all ranges except the 50 mV (10 mV/cm) range is 5 MHz at -3dB. An additional attenuator available as a separate input socket that extends the range to 300 V/cm.
The A2 amplifier is a single stage differential pair, permitting phase reversal. The gain is 300 (49.5dB) and with 5 HF compensated attenuation settings provides sensitivities from 1 V/cm to 100V/cm, with a bandwidth of 250 KHz. Clearly the A1 and A2 channels were expected to service very different needs! However, for audio and other lower frequency work, it is possible to compare two signals by superposing them.
You may have noticed that this machine has very old world connectors, designed for banana plugs and also for bare wires by means of a cross hole and a sprung shroud. They are very nice but not practical, at least in my lab, so I made a BNC adaptor plate for the A1 & A2 channels:
X Amplifier. The X amplifier provides symmetrical deflection and shift by means of a pentode differential pair. To minimise trace distortion, the anodes are DC coupled to the X plates via 3 Hivac neons in each phase that bring the plates close to ground potential. Shunt capacitors are included because the impedance of the neons rises with frequency; a series resistor is placed in each neon chain to prevent relaxation oscillation. The bandwidth of the X amplifier is -3dB at 200kHz. HF compensation is provided on the coupling network to the timebase. The gain is variable from the front panel by the provision of a variable resistance between the cathode resistors that are returned to -300V rail. The variable gain makes it possible to greatly expand an event without changing the time calibration because the time vernier-shift is applied to the input grid of the amplifier.
Synch Phase, Amplifier and Limiter and TV Synch Separator. The synch signal is picked off a tap in the anode load of the final stage of the A1 amplifier. This feeds a cathodyne stage that permits the user to select synch/trigger off positive going or negative going signals. The next stage uses a pentode that is run with its screen to a low voltage to give a short grid base. No bias is provided and so the stage will clip negative going signals at the grid and not pass positive going signals at the grid producing a clipped positive going pulse at the plate (the stage inverts) that is further amplified and inverted once again to produce a negative going pulse used to synchronise or trigger the timebase, depending on the timebase stability setting. (This also has the effect of removing positive going video content from composite TV signals. Along with this, a TV frame position of the synch control is available that passes the signal though a circuit that discriminates TV frame synch signals such that only the TV frame synch signals are passed to the timebase. I would think that this scope then, was at least in part, aimed at TV development.)
TimeBase. Written with much help from Parr and Davies, Refer to the diagram below:
The operation is best considered by considering it in the triggerable condition. During the quiescent or ‘top’ period, the suppressor grid of V11 (the 6F33 short suppressor base pentode) is biased beyond cut-off by the negative setting of RV18, the trig / free-run control. In this state, the cathode current flows to the screen grid. The plate potential which would normally rise to 350V (since plate current is cut-off by the suppressor) is clamped by the potential of the diode, V10A which is connected to the cathode of the cathode follower, V9A. (The actual value of this potential is controlled by the front panel timebase frequency control, more on this further on….) V9A presents a high impedance to the trigger signal that is applied to its grid and a low impedance to the cathode of the diode, V10A. So, the circuit is in its ‘top’ state, awaiting the trigger to cause it to run-down.
When a negative triggering pulse is applied to the grid of V9A, this pulse appears at the plate of V10A and it is passed to the grid of V12A by way of the plate of V11. From the cathode of V12A, it is applied by the Miller capacitance (the timing capacitors) to the grid of the Miller Transitron tube, V11. (The amplitude of the pulse should not exceed the grid base of V11 or it will distort the run-down, known as over-triggering. This amplitude can be controlled by the sync control on the front panel to prevent over triggering.)
The resulting negative going potential at the grid of V11 causes the screen potential to rise and since it is connected to the suppressor grid by way of R94 also drives the suppressor grid positive allowing plate current to flow and so the tube transitions (hence transitron) from not conducting to conducting. Since the suppressor current is diverted from the screen current, the screen potential will rise further and a bootstrap action takes place. The potential of the V11 plate follows the Miller run-down and is taken off to the X amplifier by way of the cathode of the cathode follower, V12A. The fall in V11 plate potential cuts off the diode, V10A and so once the run-down has started, it will not be affected by the arrival of triggering signals during the run down.
The end of the run-down is determined when the plate of V11 is too low to maintain plate current, at this point the cathode of V12A ceases to fall, but the Miller capacitance (timing capacitor) is still discharging. Since the cathode end of the Miller capacitance cannot drop further in potential, the V11 grid end of the Miller capacitance V11 rises causing a fall in screen current which drives the screen negative and cuts off the plate current. The Plate of V11 then rises and the Miller capacitance is quickly recharged (flyback) by the grid-cathode circuit of V12A until the plate of V11 reaches the clamp potential due to the cathode potential of V9A acting via the clamp diode, V10A. The circuit is now at the ‘top’ ready to re-start.
It can be seen in the oscillogram below that once the Miller step has happened, the run-down is very linear. This is because the run-down at the plate is also “seen” by the grid via the Miller capacitance, refer to the oscillogram below. Cossor noted in their manual that the plate of V11 is returned to the +350V rail allowing a high value of plate load, thus increasing the gain in the feedback loop. They also note that the use of a cathode follower in the plate-grid circuit reduces flyback time due to the low impedance at the cathode of V11 resulting in faster recharge.
It follows that as the trig / free-run is adjusted so that the suppressor of V11 is moved positive until V11 is conducting, the circuit will free-run. The function of V10B is to prevent suppressor current by limiting how far positive the suppressor potential can rise. The 6F33 and CV2209 have an internal diode for this purpose and Cossor included V10B to “assist” the internal diode probably because there was a second otherwise unused diode available in theEB91 envelope!
A very novel feature of this design is that the free-run repetition rate is controlled by the ‘top’ voltage which is set by the cathode potential of V9A. The DC potential of the grid of V9A can be controlled from the front panel using the timebase frequency control and as the ‘top’ voltage is reduced, the run-down potential change is also reduced resulting in the timebase repeating at a faster rate. Note, the actual run-down rate is fixed and determines the timing calibration. So if the operator increases the timebase frequency, he will be rewarded with a shorter timebase, to the point where it actually stops! I did not understand this when I was a kid and I have read somebody querying it on one of the forums. It appears and perhaps is, wierd. However, the actual length of the oscillogram can be increased using the X amplifier control on the front panel, as described above and so it is helpful in obtaining one or two cycle oscillograms that occupy the entire screen which is useful for photography and single stroke working.
Here is the circuit as promised:
Here are oscillograms of this timebase taken on a Tektronix 547A:
The sweep rate is 2mS/cm and from the top we have:
100 V/cm, Run-down at the plate of V11, you can see the Miller Step at the start of the run-down.
10 V/cm, V11 grid waveform, you can see the negative bump due to the trigger pulse and the run-up which is the negative feedback that causes excellent linearity.
100 V/cm, V11 suppressor waveform (used for flyback suppression or trig bright-up)
20 V/cm, Trigger pulse at the cathode of V10A, run-down starts at the negative going edge, you can see how the run-down is unaffected by further trigger pulses during the run-down.
Flyback Suppression & Trig Bright-Up. The square pulse that is available at the screen grid (see the oscillogram above) of the transitron tube is negative going during flyback and when applied to the grid of the CRT causes suppression of the trace during flyback. The screen grid pulse is also square during the sweep and positive going. By applying this positive going sweep pulse to the cathode of the CRT, it can be used to unblank the CRT during the sweep, my father termed this “trig bright-up”. On this scope, it is possible to readily change it to blank/unblank by means of a link on a side panel and it works very nicely. The advantage of this mode is that there will never be bright stationary dots (that can burn the screen) when the timebase is not running as is the case in triggered mode when there is no signal present on the A1 channel. The disadvantage is the AC coupling time-constant, which will cause the trace to dim towards the end of the sweep at lower sweep speeds.* I have found that when working above 200 Hz or so, it is not a problem so I have my unit in trig bright-up mode.
* This is the reason Tektronix developed DC coupled unblanking, I think this was done by John Kobbe. I have come across many very ingenious alternative techniques, developed by other manufacturers who were seeking to avoid paying license fees to Tektronix!
Timebase Delay. Timebase delay is introduced by switching in a delay circuit between the sync clipper and the timebase that generates a secondary pulse that is used to trigger the timebase. It is a mono-stable multivibrator that is tripped by a potential developed across a RC charging circuit. On receipt of a trigger pulse, the circuit commences to charge, the rate of charge being controlled by the operator by varying the resistance in the circuit using the delay control on the front panel. One the charge potential reaches the trip point, the circuit produces a sharp edged pulse that is uses to trigger the timebase. The delay can be increased up to at least 10 times the timebase speed in use. A diode is included in the grid circuit that cuts off when the circuit trips and remains cut-off until the circuit approaches the trip point, blocking any subsequent trigger pulses that will arrive as the circuit re-charges.
The delay capacitors are switched with the timebase range switch to maintain the correct relationship between timebase speed and delay time. Being mono-stable, it automatically resets each time it trips. The reset action speed limits its use to a repetition rate of less than 100 kHz. This feature enables the whole of a single pulse from a repetitive pulse signal to be measured without the Miller step affecting the measurement. For example if a pulse of 1 µS and 1000 Hz repetition rate is to be observed, put the time switch to 100 µS and the timebase in trigger mode. Set the time vernier to zero and use the delay control to place the start of the pulse at the centre of the graticule. Then use the vernier to bring the end of the pulse to the centre of the graticule, read the vernier and multiply by the time range in use, this will give the duration of the pulse.
WIKIpedia has this to say about AVO multimeters:
“AVO multimeters were almost ubiquitous in British manufacturing and service industry, research and development and higher and further education. They were also widely used by utilities, government agencies and the British armed forces. A number of special versions were produced to British Admiralty and Air Ministry specifications and for other customers. The Model 8 Marks V, 6 & 7 were designed to meet a NATO specification and were standard issue to NATO services. Many commercial and military service manuals specified that values for measurements of current or voltage had been made with a Model 7 or Model 8 Avometer. Advertisements of the late 1930s compared the utility of the Avometer to the slide rule. Even nowadays it can still be found in regular use.”
The manual states:
After repair the instrument should meet the following accuracies under reference conditions outlined in BS 89 1977 and IEC 51:
D.C. Voltage and Current Ranges: +/- 1% of f.s.d
A.C. Voltage and Current Ranges: +/- 2% of f.s.d. (50Hz)
Frequency Response: Variation from reading at 50Hz on ac current ranges or ac voltage ranges up to 300V not greater than +/- 3% between 15Hz and 15kHz.
Temeprature Effect” Variation due to temperature change not greater than 0.15%/C.
This polystyrene cased unit is a pale reflection of the older bakelite units I remember from my youth, especially the metal cased “Panclimatic” units that were made for tropical service. These had (as I remember) a brown bakelite top panel and a gorgeous brown hammer finish on the case. By this time, AVO was acquired by Thorn who (in my biased view) were the culprits in the demise of the quality of much of the British electronics industry. It also has plastic film wiring that must have reduced costs.
I encountered two problems with it, one the meter was stuck and two there was an intermittent fault that I eventually traced to a cold solder joint. The stuck meter was due to the central magnet being loose and it had shifted, jamming the coil. I carefully re-centered it and used a little dab of varnish in suitable locations to fix it in place. I did not attempt to investigate exactly why it was loose.
Here it is in the solid black leather case with the superb quality AVO test prods:
I keep opening my posts with something like “this is an interesting _____” and so it is. An instrument that is designed to measure from 10 μV to 1 V and 10 pA to 3 mA has little if any practical value to me, yet it is of interest. I had never given the issue of measuring such low electrical values any real thought and the design of this instrument is completely new to me.
I must preface the following attempt to describe the operation of this instrument by saying (as I have said before) that I am an M.E. and asked my friend Kenneth Kuhn http://www.kennethkuhn.com/hpmuseum/ to straighten me out which he kindly did and any errors are due to my failure to correctly transcribe his explanation:
HP in their manual have this to say, “If the 425A is considered a black box with two input and two output terminals, it is a DC amplifier”. If the amplifier were DC coupled, any drift or DC offsets in the amplifier would swamp the extremely small signal. The system uses lamps acting on photoresistors through a chopper wheel to modulate the incoming DC signal so as to present an AC signal to the AC coupled amplifier, the signal is then demodulated to restore the DC.
The output of the modulator is a rough squarewave having an amplitude proportional to the applied DC signal. The chop or modulation frequency is 5/6 of the line frequency to ensure that line noise is not passed, (50 Hz is not monotonically related to 60Hz.) The resulting AC signal is amplified by an amplifier that has a 50 Hz reject T filter in the negative feedback path thereby causing it to respond extremely strongly to the 50 Hz signal and not much else. This greatly amplified signal is next demodulated by a second light chopper that is synchronized to the modulator by virtue of using the diametrically opposite side of the same chopper wheel. A 50 Hz reject T filter removes any residual modulation resulting in a clean DC signal proportional to the input, that is applied via a cathode follower to the meter.
Kenneth also pointed out that that specialized op. amps having zero dc offset, known as commutating auto-zero amps, are available. What used to take a motor and chopper wheel is now packed in an integrated circuit. I suppose that is progress but not as much fun!
The modulator lamps and the light pipes to the photocells are visible above.
These instruments featured individually photo-calibrated meters and the original tag is still with this one:
I also have the proper probe for it, shown with the unit in the top picture.
The unit arrived in excellent condition. The meter glass was loose due to the retaining clips having fallen out. Actually getting the bezel off was a little tricky since it had been secured using brass break-off head screws, presumably to protect the special calibration. I managed to grasp each one using hemostats and got them all out without any unwanted adventure. I cleaned the glass (but NOT the meter face) and re-tensioned each clip and put a little glue on each also. The chopper bulbs were not burning consistently due to corrosion on the pins, another job for Deoxit. It then worked and applying a DC source set to plus and then minus 0.5000 on my HP 3468A, the meter showed precisely + and – 0.5. My next job may be to construct a precision DC voltage source using something like a GR decade box as a precision divider. It will be powered by batteries for quietness, if, if I do the job!
There are a couple of unusual considerations when using a DC meter of such high sensitivity:
Galvanic and thermoelectric effect between the probe and the terminals of interest. (I have in the past been puzzled by a DC shift on a HP 122A scope that turned out to be a galvanic issue. That one took some lateral thinking on my part to identify.) Also noted in the manual is the possibility of seeing triboelectric effects due to flexing of coaxial cables.
Here is the other side showing the rather daunting range switch:
Here is the underside also showing the range switch:
This is an interesting unit being capable of measuring the frequency response of an audio system over a range of 20Hz to 20kHz. Back in those days, 20kHz was considered enough? My ears give out at 6kHz and I can hear music very well so yes, 20kHz was enough. Maybe we have evolved so fast that our hearing has caught up with that of dogs in this peculiar age of man where technology, fast taking on a life of its own, is getting away from us and the average or even above average human cannot keep up, much less appreciate the ramifications of technological run-away. OK where was I? Oh yes, this ancient audio frequency test unit.
It contains a Wien bridge oscillator, stabilized with a lamp exactly like the original oscillator that Dave and Bill produced to start the amazing company we now know as Agilent. It has three decade frequency ranges, 20 Hz to 200Hz, 200 Hz to 2kHz and 2 kHz to 20 kHz. A push-pull power amplifier (using triode connected 6L6s) is included that allows for an output power up to 5W into loads of 50, 200, 600 and 5000 ohms. Load matching is accomplished by a switched impedance matching transformer. I find the output power useful for in my messing around, sometimes a power signal is handy without having to resort to hooking up an amp. The distortion is given as less than 1% at frequencies above 30Hz. Again, somewhere down the line when transistor amps having large amounts of negative feedback were introduced, our hearing suddenly evolved and we could hear distortion levels of 0.01% or less! Amazing really, how marketing types drive human evolution….. Or not.
There is also an accurate output attenuator, referenced to 0 dBm (1 mW into 600 ohms = 0.7745 Vrms). It has a range from 0 dB to -110 dB in 1 dB steps. A 600 ohm load that can be switched in is provided, allowing accurate setting of the output in dBm. The output is available as balanced or single ended.
To measure the frequency response of the device under test there a VTVM calibrated in dBm from -5 to 8 dBm and in volts from 0 to 2 Vrms is provided. This meter includes an accurate stepped attenuator that extends the meter range to 48dBm and 200 Vrms in 5 dB steps.
Here is the top of the chassis: The push-pull output transformer is directly below the power transformer and has very fine laminations. The tuning condenser is self-evident. To the right is the top of the impedance matching transformer and below that, the switched output attenuator. If you look closely, you can also see the oscillator stabilization lamp that is located between the left-most electrolytic and the 6J7 tube (that has a top grid connection).
It was working as received, I did the usual Deoxit job on the switches and tube pins and lubricated the geared tuning drive. The only other thing required was to re-calibrate the output meter and the input VTVM, pre-set pots are provided for both meters.
Another interesting item from HP, capable of measuring true RMS AC voltages over a range of less than 1 mV to 300 V over a frequency range of 10Hz to 4MHz.The accuracy is specified as:
+/- 1% FSD, 50Hz to 500KHz
+/- 2% FSD, 20Hz to 1MHz
+/- 3% FSD, 20Hz to 2MHz
+/- 5% FSD, 10Hz to 4MHz
My unit after repair does meet this specification, it has a large mirror scale meter that is easy to read. I especially like the old-fashioned arched appearance of the meter, almost architectural!
The input impedances are given as 10 MΩ shunted by 15pF on ranges 1 V to 300 V and 25pF on ranges 1 mV to 300 mV. It uses a 4 stage closed loop amplifier and about 51dB of negative feedback to secure good stability and repeatability, the input stage (outside the loop) is a cathode follower. All the amplifier tubes including the CF are 6CB6 pentodes. The power supply is, of course, regulated. The regulator is a typical series triode with a pentode error amplifier and 85 V gas tube reference. The design incorporates feed-forward to the screen of the error amplifier to secure good line-regulation.
All the manuals I have found are for units having a serial number prefixed with 313 while mine is unit 5641. The overall topology is the same, however there are many significant detail differences. I do not know which came first, I suspect that my unit predates the 313 prefix. So to get things started, here is a picture showing it working:
The main difference is that the attenuator resistor chain forms the load for the cathode follower while the schematic in the manual shows a separate cathode resistor with the CF being coupled to the attenuator via a 5μF capacitor. The attenuator in my unit has a separate 51nF cap for each range, I cannot see any reason why it could not have been implemented using a single capacitor though. Clearly the front end was the subject of some extended development! Here is the schematic of the attenuator/front end of my unit:
The wire boards are the precision attenuation resistors that being in series, are used as the CF load resistor.
As you may expect, I had to replace all the coupling capacitors in the attenuator. The other two capacitors that are critical and required replacement were the full-wave voltage doubler capacitors for the meter circuit. Any leakage will result in an unbalanced circuit and a residual deflection of the meter. Unbalance with the penultimate amplifier stage removed left the meter still showing an indication, replacement of the signal rectifier capacitors cured the issue. Other than that, the reservoir capacitor which is one section of a four section twistlock failed. I found a replacement on Ebay.
The main job was sorting the 6CB6 tubes according to microphony, they all have good transconductance. I did this by simply exchanging tubes with the first stage and tapping the unit to note the meter disturbance. Having found the least microphonic tube, I repeated the procedure with the rest of the tubes, still using the first stage location, until I had them ranked. I then installed them, least sensitive in the CF location, next in the first stage location and so on. At the end, I was rewarded with a unit that would show a perhaps 1/3rd scale flick instead of several wild full scale disturbances. As always, Deoxit is absolutely required on all tube pins and switch contacts. I also wrapped the CF and the first tube tightly with PTFE plumbers tape, just one width wide, I did not cover the whole tube, the heat has to radiate! Further, I similarly wrapped the first stage tube shield with PTFE tape (the first stage has the only shield) and slipped thin tubing over the conical spring (with considerable difficulty I note).
And here is the left side:
This arrived filthy inside, with two broken tubes and a perished cord.
The range is from 3Hz to 100KHz at “better than” +/- 2% accuracy. I found that it is indeed better than +/- 2%. Here it is after cleaning up, re-stuffing the two power supply reservoir electrolytics and replacing all the paper in oil caps:
Here are the original caps. One cap (the flat one) looks as though it has already been replaced but looking at a picture in the manual, it seems that all these caps are original. One of the two series connected power supply electrolytics had dried out completely, I re-stuffed them both, the cardboard sleeves make it easy to do this because they cover the results of cutting the aluminum cans open. Many of the caps had cracked cases and you can see that a chunk of the case had fallen away from one of them. I have never seen caps in this state before (and I have quite a bit of experience), this combined with the perished power cord suggests to me that this unit had spent much of its life in a harsh environment:
Here it is after repair showing the 60Hz self-test:
And here at 15KHz:
There is a resistor board that is used to correct each range thereby avoiding the need to use precision capacitors. I spent quite a bit of time working with this using a Tek 180A temperature stabilised crystal time marker as the frequency source:
I think it is worth describing the circuit action of this instrument, as I am not an E.E. I found it both novel and fascinating:
Refer to the block diagram above and the oscilloscope traces below:
The signal is amplified and applied to a conventional Schmitt trigger, the output of which is rectified and differentiated to apply a negative pulse (t1) to the A side of a multivibrator. This turns the A side of the MV off causing a positive pulse at the A plate that is applied to Phantastron run-down circuit, starting a linear run-down. At the start of the run down, the screen grid of the Phantastron rises and stays positive until the end of the run-down. This positive pulse is applied to the B grid of the MV (the constant current generator in the diagram) and is limited against a reference voltage by a clamping diode resulting in a controlled current pulse. The Phantastron run-down is stable and predictable in rate and duration and at the end, the screen voltage drops*, cutting off the current pulse (t2). This action results in a constant current pulse which is stable in duration. At the same time, degenerative action of the MV cathode resistor combined with the grid diode clamp causes the pulse to be stable in amplitude. One pulse is passed through the meter circuit for each input cycle. Since the meter has a large capacitor across it and the meter itself acts to discharge the capacitor, the more frequently the capacitor is charged, the higher the average charge voltage and the higher the indication on the meter.
The traces below were recorded on a Tek 547 with a 1A4 plug in:
Top, MV A plate.
Second down, Phantastron screen pulse.
Third down, Phantastron run-down.
Bottom, MV B current pulse.
I Believe the Phantastron is one of a number of extremely clever devices invented by Alan Blumlien just before or early in WW2 and so named because what it could do was, at that time, fantastic. Such triggereable timing and logic elements included multivibrators and were essential elements of radar and of course code breaking computers.
*It is fairly easy to see how the run-down commences, when a positive pulse is received at the screen grid, the tube starts to conduct, discharging the timing capacitor. The run-down termination process is not so obvious: The level at which the discharge ceases is determined when the potential between the anode and the suppressor grid becomes so low that the anode cannot draw through the suppressor grid the electrons that have passed the screen grid and so the remaining electrons return to the screen grid causing the potential of the screen grid to suddenly drop and so the tube cuts off ending the discharge.
Here is an interesting device that can be used to check oscilloscope Y rise time and trigger jitter. It came with a bunch of Tektronix type 547 oscilloscope parts.
It will change state extremely rapidly, entering a negative resistance region where as the forward current is increased the current suddenly drops causing a sudden voltage increase at the anode. It can be used to build extremely fast negative transconductance oscillators and in the trigger circuits of fast oscilloscopes. The static forward resistance of the diode is just a few ohms and in this case, this small resistance forms the shunt leg of a potential divider, since the input resistance is 3.48k, there is almost no output until the diode changes state. In this case, the change in state occurs at an input threshold of 32.5V. Here it is driven by a sine wave shown on a type 545A oscilloscope:
Here it is on the type 545A at 200nS/cm driven by the square wave calibrator set at 50V P-P. The comparative slowness of the leading edge of the square wave to that of the tunnel diode is very clear!
The slight forward lean just visible at 200nS/cm on the edge of the tunnel diode state change, maybe 15nS, is mostly due to the rise time of the type 545A Y amplifier which is itself, extremely quick. This is why for many years, tunnel diodes were used to generate the trigger pulse for fast oscilloscopes being at least an order of magnitude faster than a triode or transistor Schmitt trigger. Type 545A pre-dated the use of tunnel diodes for triggering and at high frequencies, the timebase is synchronised, not triggered. The later type 547 used tunnel diode triggering allowing the timebase to be triggered at any speed the Y amp could handle (in the case of type 547, greater than 50MHz). Here is the same set up viewed on a type 547 oscilloscope:
Interestingly, the trace on the type 547 is less clear than that of the older type 545A, I think the CRT in this type 547 is getting near the end of its life. Also, disappointingly, I could not get a jitter free trace at a speed higher than 200nS/cm. The type 545A is proving its mettle!
In an attempt to do better, I tried on a type 475 and the result was worse, I could not obtain a jitter free trace above 1µS/cm!
This type 475 is a real disappointment being a 200MHz scope. The triggering is weak and the trace definition poor. I cleaned all the switch contacts and that yielded an improvement, the trace went from very fuzzy to fuzzy and the Y stability improved. However, when I want to see something tricky, the 547, 545A or even 535A and 535 types perform better. If you are expert in restoring the performance of a type 475, or know somebody who is, please leave a comment so that I can contact you!