(Please click on this picture to view it properly. I think that the meter on the 190 A is better looking than the meter on the 190 B.)
I acquired a clean 190 B sans attenuator which is essential to the function of this design. One of my friends at VintageTek said that they had a unit on the scrap pile that had retained its attenuator so in due course that unit, which turned out to be a 190 A, arrived here. Even though it was dirty and missing the covers, one tube was missing and another tube had a failed heater (which is rare in my experience) I could not resist checking it out. In addition, two of the PSU capacitors were dry and I initially reformed them, then later, replaced them. Having put in the missing 6C4 oscillator tube and replaced the dead 12AU7, being Tektronix of course it worked, and worked properly too. The unit also had a 4 pin Jones socket mounted on the back, presumably so as to use it as a regulated power supply? I could not tell for sure since the socket had been disconnected.
As luck would have it, an attenuator turned up on Ebay for a reasonable price so I now have a working 190 A and B! As always, I applied Deoxit to all tube pins and switch contacts.
The basis of the unit is a Colpitts oscillator with 5 switched ranges covering 350KHz to 50MHz which at the time, was sufficient to support bandwidth testing of oscilloscopes. There is also a fixed 50KHz output. An output attenuator is provided at the end of a lead that has a male UHF connector mounted on it for direct connection to an oscilloscope. The attenuator is coupled to the generator by a special purpose Cannon connector that has a VHF coaxial connector enclosed with 3 other pins that support feedback of a DC signal that is linearly related to the peak-to-peak amplitude of the HF signal at the attenuator. The attenuator has 7 ranges from 0.1 to 10Vp-p, (a constant variation control is provided on the generator). The attenuator contains diodes that sample and rectify the HF output, resulting in a negative DC voltage that is close to the peak to peak amplitude of the HF signal, and that is linearly proportional to the HF signal. This DC signal is returned to the generator via the special Cannon connecter, and is used to control a regulated power supply that feeds the plate of the Colpitts oscillator and by this means, maintains a constant amplitude at the point of application, the attenuator output. The manual states that if the shunt capacitance at the output is less than 50pF, the output amplitude will vary less than +/- 2% from 50KHz to 30MHz and less than +/- 5% from 30MHz to 50MHz. Using my 200MHz Tek 475, both units appear to be very much flatter than specified. The output impedance is 52 Ohms. Here is the special connector:
The DC sample voltage is also used to drive a calibrated meter that indicates the output amplitude in p-p volts, it also shows when the generator is being operated within its controlled amplitude envelope. The frequency is indicated on a vertical drum with a separate scale for each and an vertical illuminated cursor line in a window on the front panel.
In earlier equipment, the sampling diodes were a dual diode 6110 tube, later replaced in the B version by 1N87 silicon diodes. Both my attenuators have the 6110 tube since the B unit was divorced from its original attenuator. If you are looking closely, you may have noticed an empty hole near the top of the chassis; the A version had a 6AL5 double diode in this location. The meter is connected across the cathodes of a dual triode, one grid of which is connected to the sampling diodes, and the other that was connected to the 6AL5 that is connected in the same configuration as the sampling diodes. The intention was to minimise thermal drift of the thermionic diodes being registered by the meter. I suppose that I could replace the 6110 in the attenuator that I have connected to the B unit with 1N87s but it is more likely that I will instal the 6AL5 compensation circuit instead, if I do anything. Interestingly and fortuitously, the B unit retains the heater supply to the attenuator.
Here is a Wireless World advert for this instrument that I stole ages ago from another website. I can’t remember where I got it from so if it was you, please let me know and I will acknowledge you for it:
The ebay seller originally had this up at $150 which was very high. He also said that he had turned it on and some smoke came out that he didn’t think was serious. Hmmm. So I contacted him and let him know not to just turn on old equipment because damage may well result, up to and including power transformer failure*. (This author knows, he has learnt the hard way.) I later determined that the smoke had come from the negative rail smoothing resistor that was running into a dry capacitor that was a near dead short. The result was that he re-listed it at $75 with free shipping. I was the only bidder so another relic arrived on my doorstep. I must say, the seller was very prompt in getting this to me so I left him positive feedback.
*Notably, the three power transformer failures that I have precipitated were all in USM oscilloscopes (I told you that I know, damn it). My thoughts on this are that 1/ the USM scopes tend to be rather tightly enclosed, presumably to reduce RFI emissions and 2/ this, probably combined with much duty, resulted in prolonged high temperatures hence degraded power transformer insulation combined with failed capacitors. Also, my experience prior to looking at USM scopes was almost entirely with Tektronix. In their early years, Tektronix experienced a spate of power transformer failures that they addressed by bringing in Gordon Sloat to set up their own transformer manufacturing facility. Tektronix also include 10Ω carbon fusing resistors in each rectifier circuit to protect the transformer and this excellent practice may have been inspired by the intention to make not only the best performing, but the most reliable oscilloscopes possible at the time, so I was spoilt. Encountering the very interesting USM scopes certainly increased my knowledge and experience base!
The general condition of the case is good with the exception of the CRT hood which is showing some denting and loss of the black anodising. The prop tilt stand is missing also the detector probe is missing from the accessories. As usual with new acquisitions, I opened it up, to find it in fair condition but dirty and oily due to oil migration from many leaky paper-in-oil (PIO) capacitors, I ended up replacing most of them including the HV capacitors. Most of the POI caps were superbly neat molded phenolic types, it is a shame that the sealing of the phenolic around the leadouts did not stand the test of time and temperature. All the power supply electrolytics were dry. I disconnected each one and reformed them one-by-one. Only one unit failed (the one referred to as causing the sellers non serious smoke), it did reform but the ESR was so high that it was useless. (I understand that this problem can be due to corrosion of the internal connection between the aluminium foil tails and the solder tab and/or can.) Many of the tube screen cans are exhibiting “season cracking” a phenomenon whereby deep drawn brass will transition from ductile to brittle at low temperatures so this unit apart from being hot, must have spent some time at freezing temperatures too!
At this point, the real work began for it was not happy! A hard-copy manual (that I prefer) was not available, my friend Volker Klocke has the manual in pdf form on his website at
It has early printed circuit boards. The tube sockets connect to the traces by side contacts that do not overlap or engage with the traces physically, instead relying on a solder “bridge” from each contact to the associated trace, a recipe for intermittence! I spent much time re-working these connections, this job was made more difficult by the connections on the sockets which were corroded and would not tin easily, I ended up using flux and then thoroughly cleaning each area using flux-off and a stiff brush. The other task was to test the tubes. I usually do not do this however it has been my experience that the tubes in the AN/USM scopes are often exhausted (which tends to confirm my thought about prolonged hot service). In this case, most of the tubes were strong with only two exceptions including the HV rectifier. It was apparent that somebody had gone through this example at some point, evidenced by some truly horrible soldering and melted insulation. Weller soldering gun anyone? I also found that the HV circuit had been re-wired incorrectly resulting in the unblanking multivibrator (that rides on the negative end of the HV supply to allow DC coupling to the CRT grid) not working.
Failure of PIO caps shows up in two primary ways, low power supply voltage due to a leaky bypass cap(s) or a circuit is drawing too much current due to a grid associated with a cap being high. In this case the HV voltage was less than 1000V, down from 1500V and one of the supply regulator gas tubes would extinguish when I switched in the marker generator. It is worth noting that failure of coupling caps can (and do) cause power transformer failure as well as failed electrolytic caps in the power supply!
PURPOSE and outline description
This OS-57, USM-38 oscilloscope is serial number 527 and was manufactured by the Trad Electronics Corp of New Jersey. Along with many of the vacuum tube era armed service scopes this model falls into the synchroscope category whereby the timebase can be driven by an internal trigger generator that has an output on the front panel that may in turn, be used to trigger the circuit being investigated in sync with the timebase. Applications would include radar circuits and logic circuits. Somebody tried to tell me that I was wrong in using the term synchroscope, insisting that this term refers to the well known AC power phasing device. Well yes it does, and also to the triggering/triggered oscilloscope. In fact Tektronix, in their book “Using your type 535 or type 545 oscilloscope”, refers to the term synchroscope when describing how the gate pulse from the B timebase may be used to trigger both the A timebase and the circuit under examination.
Here is the trigger and marker generator assembly:
This model uses the classic (and excellent) 3WP1 CRT with 27 tubes plus 2 hivac neons hence the 30 valve claim in the Wireless World advert above.
The timebase is of the triggered multivibrator type that is ac coupled to a push-pull deflection amplifier that is dc coupled to the X plates. The timebase has 5 ranges, 10mS/in, 1mS/in, 100µS/in, 10µS/in and 1µS/in with the X gain set for a 2.5″ sweep length and the sweep speed turned fully CW. The CRT remains cutoff until the timebase sweeps, to prevent burning of the screen by a stationary dot. A sweep expansion feature is provided that allows 9X magnification of any region of the waveform (why 9X I don’t know).
It has an internal trigger generator, rate variable from 40 to 5000 pulses/S, also a marker generator with settings at 100, 10 and 1µS.
The ac coupled Y amplifier has 5 stages including a pentode long-tailed-pair that provides push-pull deflection, ac coupled to the CRT plates. The front end is a switched attenuator that presents a constant 1M / 40pF load to the input, it has 5 ranges, 1, 3, 10, 30, 100 and 300x corresponding to sensitivity ranging from approximately 200mV/in to >50V/in. A 400nS delay line is also included. A variable up to 1Vp-p calibrated signal is provided that may be switched in; this in combination with a variable Y gain control allows on the spot calibration of the Y channel. The bandwidth of the Y amplifier is 10Hz to 6MHz. A 75Ω dummy load is available that may be plugged into the CF probe socket on the front panel to provide a standardised load to the input.
Accessories that are stored in the front panel cover include a cathode follower probe, a 10x attenuator probe, a detector probe and the aforementioned plug-in 75Ω dummy load.
Timebase. The heart of the timebase is a sweep gating multivibrator that may be adjusted from astable (free running) to monostable (triggered) using the stability control that alters the degree of negative bias applied to the multi. When the bias is sufficiently negative, the multi is held in the wait state until triggered, otherwise it will free-run. The multi provides a negative gate (rectangular pulse) to drive a simple capacitor charge sweep circuit. In the wait state, the sweep generator tube is normally conducting, that is discharging the timing capacitor. Upon receipt of the negative rectangular pulse from the gate generator, that is applied to the grid of the sweep tube, the tube cuts off, allowing the timing capacitor to charge or sweep at a rate controlled by the setting (resistance) of the variable sweep speed control. When the gate multi reverts to the wait state, the resulting positive pulse turns the sweep generator tube back on, discharging the timing capacitor (flyback).
The gate multi has a time constant that causes it to pause to allow the sweep to take place before reverting to the flyback and wait state. The gate time constant is switched with the 5 sweep rate ranges and it is also varied in tandem with the variable sweep speed control, holding the gate time constant at approximately 1/10 of the sweep time constant so that the sweep amplitude is limited to about 1/10 of the charging voltage; since the first 10% of an exponential rise/decay is substantively linear this technique results in good (surprisingly so) sweep linearity. It has the further benefit (for a given setting of the X gain) of holding the sweep length constant.
The sync signal is applied to the gating multi via a coupling diode that passes only the negative going signals that are required to trigger the multi while in the wait state. (The multi is anode triggered in case you spot the apparent contradiction between the negative going trigger and increasing negative bias locking the multi into the wait state.) In the sweep state, a positive going signal would be needed to trigger the multi so that as long as the multi is in the sweep state, it is prevented from re-triggering during the sweep by the coupling diode.
The sweep gate is also used to unblank the CRT during the sweep; the positive (inverted) going gate being capacitor coupled to the CRT grid for HF unblanking. Unblanking at low frequencies is assisted by a bistable multi (referred to as the intensity gate shaper) that rides on the negative end of the CRT supply. It is turned on and off by the unblank signal and provides the necessary square topped positive unblank pulse, directly coupled to the CRT grid.
X Amplifier. The output from the sweep generator is buffered by a cathode follower that drives the X gain control. From here, the positive going sweep signal is ac coupled to the long-tailed-pair X deflection amp. The input grid of the X deflection amp is also connected to the X shift control via a diode. The direction of the diode is such that as the sweep moves positive it disconnects, and at the end of the sweep reconnects thereby restoring the grid potential to the shift potential so that the sweep always starts from the same place, reducing jitter on the display.
Sweep Expansion. The X expansion switches in a further gain stage having a gain of approximately 9. In this mode, similarly to the grid clamp above, the start of the sweep is clamped to ground to prevent expansion jitter. The extra gain stage is arranged with a bias control that causes the stage to respond from the start of the sweep and then as the bias is increased the start point moves progressively up the ramp causing the expanded display to move along the waveform under examination. The result is that any 10% of the normal sweep can be expanded and displayed on the screen.
The timebase may be driven by the internal pulse generator or from the signal applied to the Y axis or from an external signal. There is no separate trigger circuit, the sweep gate is triggered directly by an amplified sync signal from the Y axis or from the trigger generator. The way to operate this timebase is to set the sync full ACW then bring the stability down (turning CW) until the timebase free runs, then back up again until the timebase just stops. Then bring the sync back up until the timebase triggers and locks. It may be necessary to repeat this operation if you want a different sweep speed. Triggering from the Y axis or an external signal may be selected from the rising or falling signal.
Synchronisation Amplifier. The sync amplifier consists of two gain stages and the sync selector switch that allows the user to select triggering from from a rising (+) or falling (-) edge from the Y amp or an external signal, or triggering from the internal trigger generator. The gate is triggered by a negative going signal only so the sync amp is arranged to generate a negative going signal from either a negative or positive going signal; + selection causes the sync signal to be inverted and – selection preserves the polarity of the signal.
Y Amplifier and Calibrator. The 1st stage of the Y amplifier is a pentode video amplifier that is followed by the 2nd stage cathode follower followed by a series resistor from the cathode to provide the required 1k source impedance to the delay line that is terminated with a 1k resistor shunt at the input to the 3rd stage that is also a cathode follower. The 4th stage is a second pentode video amplifier which in turn drives a push-pull pentode deflection amplifier that is ac coupled to the Y plates. The delay line drive CF is also used for sync pick-off to the sync amplifier.
In addition to the constant impedance attenuator at the front end, there is a user variable gain control located between the 1st and 2nd stages that allows the deflection to be calibrated using the internal 1Vp-p 60Hz calibration squareish wave. The calibration signal is derived using a + and – diode clipper from the power transformer.
Power Supply. There are 3 DC power supplies, positive, negative and HV (negative). The 250V B+ supply is of the choke input type and this is the first time I have encountered choke input in any equipment! It is vacuum tube rectified. A tap on one side of the B+ winding supplies a half-wave tube rectified C-R-C filtered 200V negative supply, that is used for the timebase including the gate bias and the X amplifier long-tail. The negative 1500V supply is fed from a 1100V winding that is (as is usual) a continuation of one side of the B+ winding, it ends in a winding that feeds the HV rectifier filament. The half-wave rectified HV is smoothed by a simple C-R-C filter. There is a separate heater winding for the CRT and unblanking shaper bistable multi.
Bottom view, you may spot the ceramic wirewound resistor that replaced the burnt negative rail smoothing resistor that was caused by the seller “testing” (ho hum) the unit with the failed electrolytic:
I took this one in order to get a scope-mobile which are hard to acquire since people understandably don’t want to ship them. The original model label had been removed (perhaps in shame), note the replacement label above the CRT, it sets the context for this post!
The US government issued a bid request for oscilloscopes based on the Tektronix 535 and 545 as a result of (as I understand things) a complaint from industry that Tektronix had monopolised the market. Well of course they had, their instruments were at the time, the standard bearer for oscilloscope function and quality! As a result, Hickok, Lavoie Labs and Jetronix Industries produced a large number of instruments that were blatant copies of the Tektronix instruments. It is said that imitation is the best form of flattery but these were pale copies based on what I have read and my experience of the subject Hickok 1805A. I note that not many of these copies seem to have survived, perhaps proportionally more went to the landfill than did Tektronix units, who knows?
Anyway, here it is working and in calibration after about a month on my bench. It was quite a trial, considerably more resistant to my ministrations than the genuine machines. It is now working properly and in calibration, here it is displaying a 1 mS marker and the calibrator waveform:
I read a comment regarding the Hickok 1805A on one of the forums where the author said that he thought it was subsequently produced under the Tektronix brand! That is putting the cart before the horse to put it mildly; I find it surprising that anyone who is interested in vintage electronic instruments would be unaware of the ground that Howard Vollum and his company broke. And this is not a myth, Tektronix really did develop and produce the first high-performance, reliable, triggered oscilloscope available.
This 545A copy has the Hickok 1823 Plug-In which is a copy of the Tektronix type 53/54C dual channel model which has a bandwidth of DC to 24 MHz and sensitivity from 50 mV/cm to 50 V/cm. Operating modes are A, B, A & B chopped and A & B alternate. (The later Tek type CA was the same but with the addition of A+B and A-B modes which I find to be very useful. Incidentally, the ability to sum and subtract the two channels is a major advantage of a two channel beam switch scope over a two beam scope.) Here are pictures of the 1823:
The primary patent issues were the ground breaking sweep circuit (due to Dick Ropietquet) and the distributed wide-band vertical deflection amplifier. Hickok did not contest the two issues arising from the distributed vertical amplifier patents and so the litigation focussed on six issues arising from the horizontal circuits.
Tektronix filed suit in February 1961 and the action was not settled (in their favour) until May, 1970, the final award not being settled until early 1979 at just over $4 million. This suit set a precedent whereby the US government could not longer flangrantly disregard patent ownership and that is most likely why it took so long to settle.
Here is the rather neat summary of the Tektronix horizontal circuits in question (sweep generator), cited in the litigation proceedings:
“Generally and somewhat oversimplified, the horizontal circuits in question have four components: a sweep generator, a multivibrator, a signal trigger, and a delay (or hold-off) system. The sweep generator is the heart of the circuit. Its function is to generate a linearly rising voltage which, applied to deflection plates in the cathode-ray tube, sweeps the electron beam horizontally across the tube. The multivibrator controls the sweep generator. In effect, it turns the sweep generator on and off by supplying to it one or the other of two control voltages. The signal trigger supplies to the multivibrator a pulse, by which the multivibrator is actuated. The delay or hold-off system serves to prevent the multivibrator from being triggered out of time by an incoming signal pulse and, particularly, until the conclusion of a sweep cycle.”
You may read (if you have the stamina) chapter and verse on the technical issues at http://openjurist.org/445/f2d/323/tektronix-inc-v-united-states-hickok-electrical-instrument-co
I am not going to post a lot of pictures this time, it is sufficiently similar to the 545A I posted on previously, just not as well put together. To give you some idea of what was involved, here are before and after pictures of the HV supply:
I soon discovered that it would not work properly when a dual channel plug-in was set to the Alternate mode. In this mode, the timebase sends a negative pulse at the end of each sweep to change the state of the channel switch multi, thereby turning the Y channel just displayed off and turning the alternate channel on and so on. This signal originates as a positive going pulse at the screen grid of the second stage (right hand side if you will) of the timebase gate multi (which reverts to the positive state to end the sweep) which is amplified and inverted. I fooled around with this for a long time, on and off. The pulse seemed ok but in fact was of excessive amplitude and poor shape. It finally (in the way of these things) sank in that the screen grid choke may be open. The SG was still energised via the core resistor. Well, the choke was open so the SG signal was developed across the core resistor hence the poor shape and excessive amplitude. So, I got to glaring at the outside of this choke using a magnifying glass and despite absolutely no signs of abrasion or impact, spotted one end of the break. (Such breaks usually occur on the outer layer, fortunately.) After much further glaring, having located one end, I found the other and was able to tease each end out from the criss-cross winding sufficiently to be able to tin the ends and place a single strand of wire to bridge the break. It worked and Alternate action was now available. I placed a little varnish over the repair to fix it in place. I seem to be developing some expertise in repairing broken thin-as-a-hair choke and transformer windings! It comes with the territory I guess.
Oh, and I made a copy of the Tektronix 535A/545A operators booklet that should live in the little compartment on the top:
I bought this on impulse (on ebay), it caught my attention because it had been heavily modified with a transistor triggered sweep that I was interested in. If you read this post, you will be taken through my own journey of modifications aimed at improving the original repetitive sweep generator to be accurate enough to allow useful estimation of the applied signal frequency directly from the screen. Here is one of the seller’s pictures:
Here is the wreck that I received (minus cookie and penknife):
It was very badly packed with minimal protection. However, the construction is also extremely flimsy. You might observe the right angle clips with rubber grommets of which 4 were used to “support” the front of the CRT. It was dropped as evidenced by the chassis, bent downward by the mass of the power transformer:
I set it aside in disgust, then a nice 5EDP1 flat faced CRT came up for a fair price. It will work in place of the original 5UP1 with better sensitivity so I bought it, and that provided sufficient inspiration to take this mess on! I refused to pay for a manual for this, I am glad I did not for I was pointed to the KG-635 manual on Bama and neither the assembly or the operators manual includes schematics! In view of the damage, I decided to remove all the modifications and start by returning it to the original state as far as was possible given some missing parts, this required tracing the circuits and that took a bit of sleuthing. Having done that, I embarked on my own course of modifications. It became the fun modification project I had intended when I purchased it (rather than my usual repair, restoration and calibration activities). After all, it is a toy oscilloscope so why not have fun with it!
Both the PCBs were badly cracked and I had to repair many traces. I had a nice camera bezel from a Tek scope (that I replaced with an original bezel) that fits the front of the 5EDP1 perfectly with felt pads. Conveniently, it also fit the Knight bezel studs so I mounted it on the inside of the front panel to support the front of the new CRT. (The Knight bezel was clearly intended to ape Tektronix.) I also added support ribs either side of the flimsy front panel, a cross rib at the power transformer and braces to stiffen the support for the socket end of the CRT. I placed them the way shown because I did not want angle braces blocking access. The double bolted friction at the joints with the CRT support has proven quite effective:
From a mechanical point of view, this is a rather cheaply designed kit, however in context, it probably provided quite good value. It is AC coupled and I measured the HF -3dB point at 1.1 MHz, again quite good.
Both the X and Y deflection drivers are push-pull, using a 12BH7 for the Y axis and a 12AU7 for the X axis. Both channels have pre-amplifiers, in the case of the Y axis, two cascaded 6BQ7 stages that are biased by contact potential using 10 M grid resistors and no cathode resistors, the X axis uses a single 12AU7 stage. Both axis are provided with cathode followers at the front-end. The Y axis also has a 3 decade compensated attenuator.
The sweep generator is a standard flip-flop (astable multivibrator) originally using a 6BC5 pentode for the switch and a 12AU7 for the timing capacitor charge (flyback). It is designed to run-down about 10% of the capacitor charge voltage which is one way to obtain adequate linearity, the first 10% of the exponential curve is pretty much a straight line. The small run-down amplitude is the reason for a pre-amplifier after the cathode follower in the X system. The flyback pulse is taken from the charge tube plate and amplified using a 12AU7 section, the frequency response of which results in poor flyback suppression at high sweep frequencies. The flyback suppression, which is of low amplitude and negative going at source, is amplified, clipped and inverted, then applied to the CRT cathode. The amplifier has zero bias so it will only produce a positive going output pulse.
How a Flip-Flop Sweep Generator Works:
There are two tubes, let’s say left and right. The plate of the left tube is connected to the grid of the right tube while the plate of the right tube is ac coupled to the grid of the left tube. The timing capacitor is in the cathode circuit of the right tube with the run-down resistor (usually variable) in parallel with it.
Consider that the left tube is “on”, its plate is low which turns the right tube “off” effectively disconnecting it from the timing capacitor. In this state, the timing capacitor discharges, via the run-down resistor, this is the sweep state. The run-down proceeds until the potential of the cathode of the right tube falls to the point where it turns on, recharging the capacitor. As the right tube turns on, its plate potential falls sharply and the grid of the left tube receives a negative going pulse via the coupling capacitor, turning thereby turning the left tube off, this is the flyback state. The flyback continues until the cathode of the right tube goes positive with respect to its cathode, disconnecting it once again, the turn-off causing a positive pulse at its plate that is communicated to the left hand tube, turning it on again and the circuit returns to the run-down state. The synchronisation signal is applied to the screen grid of the left tube. The pulse at the plate of the right tube is somewhat square and is generally used for the CRT flyback suppression.
While we are in this discussion, it is worth noting explicitly that though this is an oscillator, it is of the multivibrator (MV) type rather than a resonant circuit in a feedback loop. The frequency or repetition rate is given by the reciprocal of the period which is the sum of the run-down time and the flyback time. The run-down time is governed by three parameters: Value of timing capacitor, (frequency range setting) the discharge current (frequency vernier or multiplier) and the amplitude of the run-down. The flyback time also depends on the value of the timing capacitor and the amplitude however, since it is flyback, the capacitor is charging which is driven by the available recharge current.
In brief, the timing capacitor value combined with the discharge & charge current values governs the time the MV takes to go from one state to the next for a given amplitude of the waveform.
Here is the schematic of the (modified) sweep generator which may help:
1. B+ Supply:
On 120VAC, the voltages were too high, in fact the reservoir voltage was over 500 into a 450 V rated capacitor so I revised the power supply using a smaller reservoir capacitor to reduce the voltage and a choke filter,
2. HV Supply: Added a voltage doubler to the HV supply to increase the acceleration potential to around 2.1 kV. (Maximum for the 5EDP1 is 3kV.) Given the substantially higher sensitivity of the 5EDP1 over the 5UP1 this made sense, the maximum sensitivity is a useful 20mV/Div, this change also yielded an improvement in trace definition. To improve safety a little, I also replaced the metal shaft intensity pot with one having a plastic shaft,
3. B+ Regulation: There was a B+ regulator device (for the sweep and X and Y cathode followers) that sampled the HV (CRT supply) that I surmise was intended to modulate the gain of the front end to minimise sensitivity variations with HV changes. Frankly, I did not evaluate it because this scope is not an instrument, it cannot be calibrated. As is normal with cheap scopes of the tube era, the trace tended to move around the screen a bit so I replaced the 6C4 with a OA2 gas tube stabiliser*. This improved the trace stability a lot. I also had the stability and repeatability of the sweep generator repetition rate in mind since, as noted above, the repetition rate is strongly dependent on the ramp amplitude. Note, it is necessary to disconnect the heater supply at pin 4 to use an OA2 in the 6C4 location.
* The voltage at my bench bumps around quite badly, this is actually helpful because if the work I do cannot tolerate this, then speaking frankly, it is unfinished.
4. X and Y Amps: I spent a lot of time fiddling with the X and Y deflection driver supply voltages, aimed at good linearity also, similar deflection plate voltages to minimise trace distortion. In this process, I also found that increasing the cathode (tail) resistors helped, not just with linearity and balance but also bringing the plate dissipation of the tubes within limits,
5. The coupling capacitor from the Y cathode follower to the gain control was an extremely leaky 100 µF electrolytic into a 1.5 K “gain” control resulting in the trace zooming off the screen everytime I touched the control, gradually reappearing as the cap charge returned to equilibrium. I decided to eliminate this by making the gain pot the cathode follower load. The gain pot was 1.5K and I replaced it with one that measures 2.2K which is nearer to the original 2.7K load resistor. The new operating conditions resulting for the 6AB4 are fine. This is a view of the Y cathode follower side of the main PCB, also showing the new sweep frequency range switch and the new CRT front support arrangement:
6. Sweep Generator Tube Type: The tube for the sweep generator was missing (6BC5), I replaced it with a 6BH6 which has the suppressor grid brought out to the base. I found the generator seemed to work best with the suppressor grounded rather than connected to the cathode. This means that a 6BC5 cannot be plugged in directly in place of the 6BH6,
7. Test Signal: I added a diode clipper from the 10K resistor that supplies a line signal from the heater line to the sync selector. This gives a 1.1 V P-P squarish wave that followed with a 1K / 10K attenuator provides a reasonably accurate 1 V P-P test signal at the front panel.
8. Sweep Generator: The timing caps and fine control were missing due to the previous transistor sweep generator replacement. I worked up a new design with 10, 30, 100, 300, 1k, 3k, 10k, 30k and 100k frequency steps. The fine control is replaced with a variable two-terminal cascode mosfet constant current sink to control the discharge rate of the timing cap and to run-down with excellent linearity. Here is the new sweep frequency range switch:
Here is the frequency multiplier mosfet variable current source:
The idea is to obtain accurate frequency range settings, that when combined with an accurate frequency multiplier control, may be used to obtain synchronisation with minimal signal amplitude (so as to avoid pulling or pushing the sweep frequency), such that the frequency of the subject signal may be estimated directly with a fair degree of accuracy. (This is quite unlike the crude frequency controls on the general run of cheap repetitive sweep oscilloscopes.) This actually worked out after quite a bit of playing around. I arranged the control to multiply up to 4x to cover each 3 to 10 step completely. Having done this and marked a temporary scale, I set a signal generator blind to several different frequencies and used the system to estimate the signal frequency. Most of the time, I was around 13% low improving to 7% low at lower frequencies. The error was consistent which tells me that I possibly could train myself to do a lot better but still, it proves very hard to accurately esimate signal frequency directly using a repetitive sweep oscilloscope. The problem is to accurately count the number of cycles displayed and clearly, I was under-estimating. The traditional method is to use a calibrated marker generator to put blank or bright spots on the trace, however, this is not a direct method. The message is once again, if it does not have an accurate triggered sweep, it is at best, little more than a toy.
If you decide to try this out, there may be a superior current regulator approach that could be a chip, though I would make sure that the frequency response is adequate. I used the cascode mosfet because I have used it in probably dozens of other places (check out my “Audio Notes” on http://www.richardsears1.wordpress.com if you are interested), it exhibits extremely stable current control over a very wide range of impressed voltage. Also note that the mosfets will fail if you somehow break either connection to the current regulator while it is under power. I have done this several times over the years and it is irritating!
The multiplier control requires an inverse power characteristic, to a first approximation, the law of the control looks like R = 35.37X^-1.1 where R is the current regulator sensing resistor in KΩ and X is the multiplier value (in this case from 1 to 4) that ideally corresponds to uniform increments of rotation. I found a reverse log pot to be workable. The actual discharge current ranges from 72 µA to 320 µA over the multiplication range.
Lastly, I added a frequency calibration preset control on the front panel since the calibration does drift somewhat during warm up. I did not want to put it into the cabinet, only to find my efforts rendered useless due to increased temperature. This preset is also necessary to accommodate replacement of either of the sweep generator tubes. It can be set at line frequency by ensuring that 6 whole cycles are displayed on the 10 Hz setting and 2 whole cycles on the 30 Hz setting. I also added a sweep out terminal to permit the use of a frequency meter to set the sweep frequency calibration. The calibration control was implemented by replacing the 1k plate resistor of V5a with an 850R resistor and a 250R RV4 type preset pot, wired as a variable resistor in series with the 850R resistor.
Here it is working showing my crude attempt at new control legends; this is one way to display two waveforms on a single channel oscilloscope!
The schematics for the KG-630 do not seem to be on the internet so these may help if you have one of these contraptions.
INTRODUCTION. I had one of these when I was a kid. It was my first experience of a ‘proper’ oscilloscope, capable of making measurements and started my (admittedly eccentric) fascination with these machines. This was around 1974 and my dad, a Chartered Electronics Engineer and Senior Scientific Officer with the UK Met Office Cloud Physics lab, still used a Cossor 1035 MKIII at his development bench while the technicians were using Tek scopes. The original version when it was introduced, may have been the first dual-beam, triggered oscilloscope capable of making measurements. Since I wrote that, I found a June 1950 Wireless World advert for the original 1035 and from the picture I can see that it did not have the timebase delay that the MKIII has. Tektronix did not come out with a dual channel scope until the 530/535 in 1954 so Cossor did beat them to it. However, once Tek did bring the 530 series, everything else was in the shade. That it took HP 10 years to catch up (barely) really says it all for Tektronix. Anyway, here is the Cossor ad:
ARRIVAL. I had been wanting one of these and a guy in England contacted me saying he had just acquired one and was looking to sell it. The pictures showed a MKIII example in good condition so I took a chance on it and amazingly, just 5 days later, UPS arrived saying “we have a machine for you”. I unwrapped it with no little anticipation and was much relieved to find it undamaged. I think it is handsome, it is also surprisingly capable:
SPLIT BEAM CRT. It is based on what as far as I know is a unique Cossor technology, the split-beam CRT. (A. C. Cossor got into business initially in 1859 as a manufacturer of scientific glassware, later X-ray tubes then vacuum tubes, diversifying into electronics after WW1.) The tube uses single Y plates that are placed either side of a beam splitter that is in mechanical and electrical contact with the final anode. The location of the Y plates either side of the splitter largely shields each beam from ‘fringing’ or interference by the opposite Y plate. The shielding is rendered perfect at the anode end by the contact with the final anode. At the screen end, fringing is reduced by means of two ‘bucking wires’ that are cross-connected to the opposite deflector plates and serve to annul the stray fields such that the interaction between beams is reduced to less than 1%. Correction for equal beam brightness is made using a small permanent magnet near the socket end of the tube that by deflecting the beam before it meets the splitter plate can be adjusted to obtain equal beam brightness. (Source: The Cathode Ray Tube And Its Applications, Parr and Davie.)
I recall that the unit that I had as a kid exhibited the best beam definition I had ever known and this seems to be true, even the best of the Tek CRTs falls a little short and most, far short. It is simply beautiful to use.
CONDITION & REPAIR. Upon arrival, I found it to be fairly dirty inside, otherwise in excellent condition. I brushed the loose dirt away, I may wash it in the summer when there is strong sun to dry it. Then came the usually Deoxit treament of all the tube sockets. Next was to replacing the dried reservoir and smoothing caps (I disconnected them and checked them using a bench PSU). The flyback suppression coupling cap (visconol) was leaking oil so I replaced it right away. On turning it on I was rewarded with traces but a whole lot of hum showing. This turned out to be due to leaky coupling caps in the A1 amp causing heavy current draw that was pulling the B+ down hard enough to cause the regulator to drop out. Replacing those resulted in quiet traces and we were on our way! I wound up replacing most of the small paper in oil caps including the Plessey “toffee” types used for the timing of the timebase and delay circuits. They were all leaky causing extremely poor linearity:
The mains selector permits operation on 120 V so I fitted a new US style power cord. The only tube replacement required was the short suppressor base (in the US, dual-control) pentode used in the timebase. (These types have an unusually close wound suppressor grid to achieve the short base.) Cossor used a 6F33 which I replaced with a CV2209 (which I think may have been a special quality tube type for Racal). After taking care of these simple issues, I was able to calibrate it.
Here it is displaying 1 mS markers and a squarewave that is synched to the markers:
Note the great trace definition! The distortion at the start is due to a disadvantage of the transitron based timebase, the Miller step, whereby the anode voltage drops a small amount suddenly at the initiation of the sweep. I say more about this under the design notes below.
PERFORMANCE & HISTORICAL CONTEXT. It has two AC coupled channels, the A1 channel having a response of 5 Hz to 5 MHz with a maximum sensitivity of 10 mV/cm; the A2 channel has a response of 5 Hz to 250 KHz with a maximum sensitivity of 1 V/cm.
The timebase has 9 ranges, 100 mS to 10 µS. X expansion is available permitting a maximum spot velocity of 250 nS/cm. The reason the timebase ranges are stated as time not velocity is because the time ranges relate to a vernier scale on the X-shift; time is measured by setting the vernier at zero with the start of the event at a chosen location on the screen graticule, then using the vernier, sliding the end of the event to the same location. The time is given by multiplying the vernier reading by the time range is use. Voltage on the Y axis is measured using the same slide-vernier technique, the value of the event height read from the vernier is multiplied by the Y attenuator setting value in use to give the voltage. This method seems archaic but in use is very simple and easy. In fact, the timebase delay technology introduced by Tektronix handles magnified event measurement in the same way. At that time, nobody had introduced regulated CRT HV supplies and this was a good way to avoid the effect of HV voltage related CRT sensitivity variation on measurement accuracy. (Again Tektronix introduced CRT supply regulation.)
Timebase delay is also included to permit event expansion, such as the viewing and measurement of the whole of a pulse across the screen width. The specification states that it will measure time and voltage to better than 10% which nowadays in our Parts Per Million world sounds hopeless, but in those days of oscilloscopes that were not capable of calibration was good news! In my estimation I would say it is better than 5%, not far short of the new standard set by Tektronix. I don’t know when the first 1035 MKI model was released but it must have been around or maybe slightly before the ground-breaking Tektronix scopes first became available. Unlike the first Tektronix, the 511, this one has regulated power supplies, absolutely unheard of before then. Tektronix did not regulate the power supplies in their scopes until the 511A.
The timebase is a triggerable sweep duration rather than frequency design and this permits time measurement without resorting to a marker generator. It can be operated in triggered or repetitive modes. Here is the first reference that I have found to this design again, from Parr and Davies:
Martin on the british Vintage Test Gear & Workshop Equipment forum pointed out the the EAC91 specified is a single diode triode, it looks like an error and may have been intended to be EBC91, thanks Martin!
WHAT IT LOOKS LIKE INSIDE. Before I get into a more detailed technical description, here are pictures of it with the covers off, there is a lot of space due to the original design having used octal tubes!
The A1 amplifier is on the shelf while the A2 amplifier is on the deck at the front. The power supply regulators are on the deck at the back.
The timebase occupies the front section with the CRT circuit at the back.
Note the handsome potted double C core power transformer! I also like the Colvern ganged wirewound precision potentiometers used for the measurement vernier scales.
Cathode Ray Tube. The design is based on a Cossor split beam, flat face CRT, the major consequence of which is that the Y amplifiers (referred to in the manual as A1 and A2) are single ended. This might sound like a major compromise and in comparison with more modern designs I suppose it is however, in use, the amps seem to provide highly accurate, symmetrical traces. Despite the lack of common mode rejection or DC heater supplies the traces are perfectly free from visible noise. The X and Y deflection plates are available for direct connection via slide doors in the oscilloscope cabinet.
The cathode is operating at -2kV with the first and final anodes at ground potential. A Post Deflection Acceleration anode is provided, normally run at +350V with provision to increase the PDA to 2kV when extra brightness is required. All deflection plates are run at DC ground potential to minimise trace distortion. The focus is exceptional despite the lack of astigmatism correction.
Here is another picture of it showing gate and sweep waveforms from a Tek scope:
Power Supplies. The basis of the power supply is a beautiful potted C core transformer. A single 330-0-330 winding serves both the positive and the negative rails, the positive rectifier being a tube and the negative side is handled by two selenium rectifiers.
The positive side is LC smoothed to provide a +350 V rail that feeds a classic series regulator having a triode connected pentode for the series pass element and a pentode error amplifier. The regulator supplies +200 V to most of the circuits in the scope.
The negative side is RC smoothed to a -300 V rail and a gas tube stabilised -150V rail that also serves as the reference for the +200 V regulator.
The CRT supplies are taken from a single HV winding and recified using selenium sticks both ways to provide plus and minus 2 kV, these supplies are RC smoothed.
Making Measurements. At this point, it is appropriate to further expound on the vernier-shift measurement technique used for both time and voltage measurements on this oscilloscope. Here is a diagram showing the electrical arrangement:
The method of measurement is; set the cursor of the appropriate vernier to zero on the scale on the panel, then use the shift potentiometer to set the lower (voltage measurements) or left (time measurements) part of the event to a chosen location on the graticule, usually the centre. Then operate the vernier until the other end of the event is at the chosen location. Read the value on vernier scale under the cursor and then multiply by the voltage or time range setting to give the voltage or time value.
The two variable resistors RV13 & RV15 are a ganged pair that comprise the shift control, the wiper of RV13 is connected to the vernier potentiometer RV14, while the wiper of RV15 is connected to the other end of the vernier potentiometer, the shift potential being taken from the wiper of RV14. RV13 & RV15 have no scale, they allow the beam to be shifted by moving the entire vernier pot up and down electrically whilst maintaining a constant potential across the vernier pot and thus, the calibration of the vernier scale. The vernier-shift network is fed from regulated supplies to maintain calibration.
The calibration is accomplished using RV12 whereby changing the setting of RV12 will change the potential across the network and thus the vernier pot, changing the amount the beam moves for a given movement of the vernier pot. RV12 is manipulated until the vernier scale accurately corresponds with a known voltage or time being displayed. For example, if the attenuator is set at 5 and a 5 V p-p event is on the screen, with the upper peak on the centre and the cursor at zero, then shifting it downward from zero using the vernier until the lower peak is at the centre should place the cursor over 5 on the scale.
The graticule is engraved on both sides that in combination with the flat screen makes it easy to avoid parallax error. It should be noted that due to the Miller Step, Cossor recommended avoiding the first 10% of the trace when making time measurements. Physically, the vernier pot is mounted at the front of the shift pair and operated by a spindle that runs coaxially around the spindle of the shift pair control spindle and has a nice hairline cursor knob over a vernier scale that is printed on the front panel. The spindle of the shift pair emerges at a simple small control knob in the centre of the vernier.
Here is what the controls look like:
The beautiful Colvern wirewound pots used for these controls are sufficiently linear that the use of a printed scale works very well. This is in marked contrast to the silly printed scales on the front panel of toy oscilloscopes!
Y Amplifiers. The A1 amplifier is the primary Y channel and the trigger / synch signal is taken from this channel. One aspect that is odd is that the A1 deflection is positive down, I would love to know the design rationale for this arrangement…….. This could have been overcome by mounting the CRT the other way up! The A2 amplifier is a simple differential pair and phase reversal is provided on the range switch so if the CRT had been mounted the other way up, it would have been simple to have the panel phase marking for the A2 channel correct. (I am actually considering this modification, if the CRT wiring will permit. All that is needed to correct the A2 channel (to maintain the correct front panel phase markings) is to reverse the grid connections from the attenuator to the opposite tubes!
Both channels are AC coupled which is a significant downside, probably necessary to avoid DC drift in the single ended amplifiers that would be hard to eliminate despite the use of regulated power supplies.
The A1 amplifier has a maximum gain of 3000 (69.5dB). It uses three RC connected pentodes and what the manual describes as “heavy negative feedback”. Each stage has adjustable bias, the bias and all plates and screens are fed from the regulated supplies except the final stage plate which is fed from the LC smoothed 350V supply. Seven switched HF compensated attenuation levels provide sensitivities from 10 mV/cm to 10 V/cm. The frequency response on all ranges except the 50 mV (10 mV/cm) range is 5 MHz at -3dB. An additional attenuator available as a separate input socket that extends the range to 300 V/cm.
The A2 amplifier is a single stage differential pair, permitting phase reversal. The gain is 300 (49.5dB) and with 5 HF compensated attenuation settings provides sensitivities from 1 V/cm to 100V/cm, with a bandwidth of 250 KHz. Clearly the A1 and A2 channels were expected to service very different needs! However, for audio and other lower frequency work, it is possible to compare two signals by superposing them.
You may have noticed that this machine has very old world connectors, designed for banana plugs and also for bare wires by means of a cross hole and a sprung shroud. They are very nice but not practical, at least in my lab, so I made a BNC adaptor plate for the A1 & A2 channels:
X Amplifier. The X amplifier provides symmetrical deflection and shift by means of a pentode differential pair. To minimise trace distortion, the anodes are DC coupled to the X plates via 3 Hivac neons in each phase that bring the plates close to ground potential. Shunt capacitors are included because the impedance of the neons rises with frequency; a series resistor is placed in each neon chain to prevent relaxation oscillation. The bandwidth of the X amplifier is -3dB at 200kHz. HF compensation is provided on the coupling network to the timebase. The gain is variable from the front panel by the provision of a variable resistance between the cathode resistors that are returned to -300V rail. The variable gain makes it possible to greatly expand an event without changing the time calibration because the time vernier-shift is applied to the input grid of the amplifier.
Synch Phase, Amplifier and Limiter and TV Synch Separator. The synch signal is picked off a tap in the anode load of the final stage of the A1 amplifier. This feeds a cathodyne stage that permits the user to select synch/trigger off positive going or negative going signals. The next stage uses a pentode that is run with its screen to a low voltage to give a short grid base. No bias is provided and so the stage will clip negative going signals at the grid and not pass positive going signals at the grid producing a clipped positive going pulse at the plate (the stage inverts) that is further amplified and inverted once again to produce a negative going pulse used to synchronise or trigger the timebase, depending on the timebase stability setting. (This also has the effect of removing positive going video content from composite TV signals. Along with this, a TV frame position of the synch control is available that passes the signal though a circuit that discriminates TV frame synch signals such that only the TV frame synch signals are passed to the timebase. I would think that this scope then, was at least in part, aimed at TV development.)
TimeBase. Written with much help from Parr and Davies, Refer to the diagram below:
The operation is best considered by considering it in the triggerable condition. During the quiescent or ‘top’ period, the suppressor grid of V11 (the 6F33 short suppressor base pentode) is biased beyond cut-off by the negative setting of RV18, the trig / free-run control. In this state, the cathode current flows to the screen grid. The plate potential which would normally rise to 350V (since plate current is cut-off by the suppressor) is clamped by the potential of the diode, V10A which is connected to the cathode of the cathode follower, V9A. (The actual value of this potential is controlled by the front panel timebase frequency control, more on this further on….) V9A presents a high impedance to the trigger signal that is applied to its grid and a low impedance to the cathode of the diode, V10A. So, the circuit is in its ‘top’ state, awaiting the trigger to cause it to run-down.
When a negative triggering pulse is applied to the grid of V9A, this pulse appears at the plate of V10A and it is passed to the grid of V12A by way of the plate of V11. From the cathode of V12A, it is applied by the Miller capacitance (the timing capacitors) to the grid of the Miller Transitron tube, V11. (The amplitude of the pulse should not exceed the grid base of V11 or it will distort the run-down, known as over-triggering. This amplitude can be controlled by the sync control on the front panel to prevent over triggering.)
The resulting negative going potential at the grid of V11 causes the screen potential to rise and since it is connected to the suppressor grid by way of R94 also drives the suppressor grid positive allowing plate current to flow and so the tube transitions (hence transitron) from not conducting to conducting. Since the suppressor current is diverted from the screen current, the screen potential will rise further and a bootstrap action takes place. The potential of the V11 plate follows the Miller run-down and is taken off to the X amplifier by way of the cathode of the cathode follower, V12A. The fall in V11 plate potential cuts off the diode, V10A and so once the run-down has started, it will not be affected by the arrival of triggering signals during the run down.
The end of the run-down is determined when the plate of V11 is too low to maintain plate current, at this point the cathode of V12A ceases to fall, but the Miller capacitance (timing capacitor) is still discharging. Since the cathode end of the Miller capacitance cannot drop further in potential, the V11 grid end of the Miller capacitance V11 rises causing a fall in screen current which drives the screen negative and cuts off the plate current. The Plate of V11 then rises and the Miller capacitance is quickly recharged (flyback) by the grid-cathode circuit of V12A until the plate of V11 reaches the clamp potential due to the cathode potential of V9A acting via the clamp diode, V10A. The circuit is now at the ‘top’ ready to re-start.
It can be seen in the oscillogram below that once the Miller step has happened, the run-down is very linear. This is because the run-down at the plate is also “seen” by the grid via the Miller capacitance, refer to the oscillogram below. Cossor noted in their manual that the plate of V11 is returned to the +350V rail allowing a high value of plate load, thus increasing the gain in the feedback loop. They also note that the use of a cathode follower in the plate-grid circuit reduces flyback time due to the low impedance at the cathode of V11 resulting in faster recharge.
It follows that as the trig / free-run is adjusted so that the suppressor of V11 is moved positive until V11 is conducting, the circuit will free-run. The function of V10B is to prevent suppressor current by limiting how far positive the suppressor potential can rise. The 6F33 and CV2209 have an internal diode for this purpose and Cossor included V10B to “assist” the internal diode probably because there was a second otherwise unused diode available in theEB91 envelope!
A very novel feature of this design is that the free-run repetition rate is controlled by the ‘top’ voltage which is set by the cathode potential of V9A. The DC potential of the grid of V9A can be controlled from the front panel using the timebase frequency control and as the ‘top’ voltage is reduced, the run-down potential change is also reduced resulting in the timebase repeating at a faster rate. Note, the actual run-down rate is fixed and determines the timing calibration. So if the operator increases the timebase frequency, he will be rewarded with a shorter timebase, to the point where it actually stops! I did not understand this when I was a kid and I have read somebody querying it on one of the forums. It appears and perhaps is, wierd. However, the actual length of the oscillogram can be increased using the X amplifier control on the front panel, as described above and so it is helpful in obtaining one or two cycle oscillograms that occupy the entire screen which is useful for photography and single stroke working.
Here is the circuit as promised:
Here are oscillograms of this timebase taken on a Tektronix 547A:
The sweep rate is 2mS/cm and from the top we have:
100 V/cm, Run-down at the plate of V11, you can see the Miller Step at the start of the run-down.
10 V/cm, V11 grid waveform, you can see the negative bump due to the trigger pulse and the run-up which is the negative feedback that causes excellent linearity.
100 V/cm, V11 suppressor waveform (used for flyback suppression or trig bright-up)
20 V/cm, Trigger pulse at the cathode of V10A, run-down starts at the negative going edge, you can see how the run-down is unaffected by further trigger pulses during the run-down.
Flyback Suppression & Trig Bright-Up. The square pulse that is available at the screen grid (see the oscillogram above) of the transitron tube is negative going during flyback and when applied to the grid of the CRT causes suppression of the trace during flyback. The screen grid pulse is also square during the sweep and positive going. By applying this positive going sweep pulse to the cathode of the CRT, it can be used to unblank the CRT during the sweep, my father termed this “trig bright-up”. On this scope, it is possible to readily change it to blank/unblank by means of a link on a side panel and it works very nicely. The advantage of this mode is that there will never be bright stationary dots (that can burn the screen) when the timebase is not running as is the case in triggered mode when there is no signal present on the A1 channel. The disadvantage is the AC coupling time-constant, which will cause the trace to dim towards the end of the sweep at lower sweep speeds.* I have found that when working above 200 Hz or so, it is not a problem so I have my unit in trig bright-up mode.
* This is the reason Tektronix developed DC coupled unblanking, I think this was done by John Kobbe. I have come across many very ingenious alternative techniques, developed by other manufacturers who were seeking to avoid paying license fees to Tektronix!
Timebase Delay. Timebase delay is introduced by switching in a delay circuit between the sync clipper and the timebase that generates a secondary pulse that is used to trigger the timebase. It is a mono-stable multivibrator that is tripped by a potential developed across a RC charging circuit. On receipt of a trigger pulse, the circuit commences to charge, the rate of charge being controlled by the operator by varying the resistance in the circuit using the delay control on the front panel. One the charge potential reaches the trip point, the circuit produces a sharp edged pulse that is uses to trigger the timebase. The delay can be increased up to at least 10 times the timebase speed in use. A diode is included in the grid circuit that cuts off when the circuit trips and remains cut-off until the circuit approaches the trip point, blocking any subsequent trigger pulses that will arrive as the circuit re-charges.
The delay capacitors are switched with the timebase range switch to maintain the correct relationship between timebase speed and delay time. Being mono-stable, it automatically resets each time it trips. The reset action speed limits its use to a repetition rate of less than 100 kHz. This feature enables the whole of a single pulse from a repetitive pulse signal to be measured without the Miller step affecting the measurement. For example if a pulse of 1 µS and 1000 Hz repetition rate is to be observed, put the time switch to 100 µS and the timebase in trigger mode. Set the time vernier to zero and use the delay control to place the start of the pulse at the centre of the graticule. Then use the vernier to bring the end of the pulse to the centre of the graticule, read the vernier and multiply by the time range in use, this will give the duration of the pulse.
WIKIpedia has this to say about AVO multimeters:
“AVO multimeters were almost ubiquitous in British manufacturing and service industry, research and development and higher and further education. They were also widely used by utilities, government agencies and the British armed forces. A number of special versions were produced to British Admiralty and Air Ministry specifications and for other customers. The Model 8 Marks V, 6 & 7 were designed to meet a NATO specification and were standard issue to NATO services. Many commercial and military service manuals specified that values for measurements of current or voltage had been made with a Model 7 or Model 8 Avometer. Advertisements of the late 1930s compared the utility of the Avometer to the slide rule. Even nowadays it can still be found in regular use.”
The manual states:
After repair the instrument should meet the following accuracies under reference conditions outlined in BS 89 1977 and IEC 51:
D.C. Voltage and Current Ranges: +/- 1% of f.s.d
A.C. Voltage and Current Ranges: +/- 2% of f.s.d. (50Hz)
Frequency Response: Variation from reading at 50Hz on ac current ranges or ac voltage ranges up to 300V not greater than +/- 3% between 15Hz and 15kHz.
Temeprature Effect” Variation due to temperature change not greater than 0.15%/C.
This polystyrene cased unit is a pale reflection of the older bakelite units I remember from my youth, especially the metal cased “Panclimatic” units that were made for tropical service. These had (as I remember) a brown bakelite top panel and a gorgeous brown hammer finish on the case. By this time, AVO was acquired by Thorn who (in my biased view) were the culprits in the demise of the quality of much of the British electronics industry. It also has plastic film wiring that must have reduced costs.
I encountered two problems with it, one the meter was stuck and two there was an intermittent fault that I eventually traced to a cold solder joint. The stuck meter was due to the central magnet being loose and it had shifted, jamming the coil. I carefully re-centered it and used a little dab of varnish in suitable locations to fix it in place. I did not attempt to investigate exactly why it was loose.
Here it is in the solid black leather case with the superb quality AVO test prods:
I keep opening my posts with something like “this is an interesting _____” and so it is. An instrument that is designed to measure from 10 μV to 1 V and 10 pA to 3 mA has little if any practical value to me, yet it is of interest. I had never given the issue of measuring such low electrical values any real thought and the design of this instrument is completely new to me.
I must preface the following attempt to describe the operation of this instrument by saying (as I have said before) that I am an M.E. and asked my friend Kenneth Kuhn http://www.kennethkuhn.com/hpmuseum/ to straighten me out which he kindly did and any errors are due to my failure to correctly transcribe his explanation:
HP in their manual have this to say, “If the 425A is considered a black box with two input and two output terminals, it is a DC amplifier”. If the amplifier were DC coupled, any drift or DC offsets in the amplifier would swamp the extremely small signal. The system uses lamps acting on photoresistors through a chopper wheel to modulate the incoming DC signal so as to present an AC signal to the AC coupled amplifier, the signal is then demodulated to restore the DC.
The output of the modulator is a rough squarewave having an amplitude proportional to the applied DC signal. The chop or modulation frequency is 5/6 of the line frequency to ensure that line noise is not passed, (50 Hz is not monotonically related to 60Hz.) The resulting AC signal is amplified by an amplifier that has a 50 Hz reject T filter in the negative feedback path thereby causing it to respond extremely strongly to the 50 Hz signal and not much else. This greatly amplified signal is next demodulated by a second light chopper that is synchronized to the modulator by virtue of using the diametrically opposite side of the same chopper wheel. A 50 Hz reject T filter removes any residual modulation resulting in a clean DC signal proportional to the input, that is applied via a cathode follower to the meter.
Kenneth also pointed out that that specialized op. amps having zero dc offset, known as commutating auto-zero amps, are available. What used to take a motor and chopper wheel is now packed in an integrated circuit. I suppose that is progress but not as much fun!
The modulator lamps and the light pipes to the photocells are visible above.
These instruments featured individually photo-calibrated meters and the original tag is still with this one:
I also have the proper probe for it, shown with the unit in the top picture.
The unit arrived in excellent condition. The meter glass was loose due to the retaining clips having fallen out. Actually getting the bezel off was a little tricky since it had been secured using brass break-off head screws, presumably to protect the special calibration. I managed to grasp each one using hemostats and got them all out without any unwanted adventure. I cleaned the glass (but NOT the meter face) and re-tensioned each clip and put a little glue on each also. The chopper bulbs were not burning consistently due to corrosion on the pins, another job for Deoxit. It then worked and applying a DC source set to plus and then minus 0.5000 on my HP 3468A, the meter showed precisely + and – 0.5. My next job may be to construct a precision DC voltage source using something like a GR decade box as a precision divider. It will be powered by batteries for quietness, if, if I do the job!
There are a couple of unusual considerations when using a DC meter of such high sensitivity:
Galvanic and thermoelectric effect between the probe and the terminals of interest. (I have in the past been puzzled by a DC shift on a HP 122A scope that turned out to be a galvanic issue. That one took some lateral thinking on my part to identify.) Also noted in the manual is the possibility of seeing triboelectric effects due to flexing of coaxial cables.
Here is the other side showing the rather daunting range switch:
Here is the underside also showing the range switch: